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Experimental demonstration of techniques to improve system performance in non-static microwave frequency analog and digital signal transmission over fiber -optic communication systems

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Experimental demonstration of techniques to improve system performance in non-static microwave frequency analog and digital signal transmission over fiber -optic communication systems

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Content EXPERIMENTAL DEMONSTRATION OF TECHNIQUES TO IMPROVE SYSTEM PERFORMANCE IN NON-STATIC MICROWAVE FREQUENCY ANALOG AND DIGITAL SIGNAL TRANSMISSION OVER FIBER-OPTIC COMMUNICATION SYSTEMS by Asaf Sahin A Dissertation Presented to the FACULTY OF THE GRADUATE SCHOOL UNIVERSITY OF SOUTHERN CALIFORNIA In Partial Fulfillment of the Requirements for the Degree DOCTOR OF PHILOSOPHY (ELECTRICAL ENGINEERING) August 2003 Copyright 2003 Asaf Sahin Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. UMI Number: 3116779 INFORMATION TO USERS The quality of this reproduction is dependent upon the quality of the copy submitted. Broken or indistinct print, colored or poor quality illustrations and photographs, print bleed-through, substandard margins, and improper alignment can adversely affect reproduction. In the unlikely event that the author did not send a complete manuscript and there are missing pages, these will be noted. Also, if unauthorized copyright material had to be removed, a note will indicate the deletion. ® UMI UMI Microform 3116779 Copyright 2004 by ProQuest Information and Learning Company. All rights reserved. This microform edition is protected against unauthorized copying under Title 17, United States Code. ProQuest Information and Learning Company 300 North Zeeb Road P.O. Box 1346 Ann Arbor, Ml 48106-1346 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. UNIVERSITY OF SOUTHERN CALIFORNIA THE GRADUATE SCHOOL UNIVERSITY PARK LOS ANGELES, CALIFORNIA 90089-1695 This dissertation, written by 8 . SAUTl\i _____________________________ under the direction o f h t 5 dissertation committee, and approved by all its members, has been presented to and accepted by the Director o f Graduate and Professional Programs, in partial fulfillment o f the requirements fo r the degree of DOCTOR OF PHILOSOPHY Director Date A u g u st 12 f 2 0 0 3 7 Dissertation Committee llo iifp r LLflRDi DANIEL iZ£CM Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Dedication To my mother and brother for their trust and support. And to the memory of my dear father. Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Acknowledgements I would like to thank my advisor and dissertation committee chairman Prof. Alan E. Willner, for his invaluable attention, guidance, and support in my graduate studies. Furthermore, I wish to thank Prof. John D. O’Brien, Prof. Alexander Sawchuk, Prof. Daniel Rich, and Prof. Robert Gagliardi for serving on my dissertation and qualifying committee. I would like to thank my colleagues and friends from OCLab. My whole endeavor would have been impossible without their knowledge and help. First the ones, who have already made it: Dr. Mustafa C. Cardakli, Dr. Steve A. Havstad, Dr. Olaf H. Adamczyk, Dr. Reza Khosravani, Dr. Sangeon Lee, Dr. Yong Xie. My generation, crop of 2003: Dr. Zhongqi Pan, Yon-Won Song, Deniz Gurkan. And my professors at Middle East Technical University (METU), who have taught and inspired me to not to loose hope. Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Table o f Contents Dedication _____________________________________________________ ii Acknowledgements_______________________________________________ iii List of Figures ___________________ vii Abstract xii Chapter 1 1 Introduction and background_________________________________________ 1 1.1 Non-static fiber-optic communication links __________________ 3 1.2 Subcarrier multiplexing in optical communication systems___________ 5 1.3 System issues in analog and digital fiber-optic links_____________ 8 1.3.1 Subcarrier Based All Optical Switching___________________ 9 1.3.2 RF fading____________________________________________ 10 1.3.3 Nonideal fiber characteristics_____________________________ 12 1.3.4 Chromatic dispersion___________________________________ 12 1.3.5 Nonlinearities 15 Chapter 2 ___________________________________________________ 18 Inline Dispersion Slope Monitoring of Many WDM Channels Using Dispersion-Induced RF Clock Regeneration_________________ 18 2.1 Introduction________________________________ 18 2.2 Concept for Dispersion Slope Monitoring Using Dispersion-Induced RF Clock Regeneration __________________________________ 20 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 2.3 Experimental Setup_________________________________________ 22 2.4 Results__________________________________________________ 24 2.5 Conclusion 25 Chapter 3 _________________________________________________ 26 Distance-independent RF fading compensation using a tunable nonlinearly- chirped FBG in a phase diversity configuration______________ 26 3.1 Introduction_________ 26 3.2 Concept of RF fading compensation using phase diversity__________ 28 3.3 Experimental setup of the compensation module ____________ 31 3.3.1 Nonlinearly-chirped fiber Bragg grating_____________________ 34 3.4 Experimental results________________________________________ 40 3.5 Conclusion 45 Chapter 4 46 Wavelength Conversion of Subcarrier Channels using Difference Frequency Generation in a PPLN Waveguide^________________________ 46 4.1 Introduction______________________________________________46 4.2 Experimental Setup_______________________ 47 4.3 Experimental Results ____________________________________ 50 4.4 Conclusion 54 Chapter 5 55 Statistics of PMD-induced power fading for double sideband and single sideband subcarrier-multiplexed signals_______________________ 55 5.1 Introduction______________________________________________ 55 5.2 Experimental Setup ______________________________________ 56 5.3 Experimental Results_______________________________________ 58 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 5.4 Conclusion 61 Chapter 6 62 Dispersion Division Multiplexing for In-Band Subcarrier-Header-Based All- Optical Packet Switching_________________________________62 6.1 Introduction_______________________________________________62 6.2 Concept^______________ 64 6.3 Experimental Setup ________ 65 6.4 Experimental Results___________ 68 6.4.1 Receiver RF Spectra______________________________________ 68 6.4.2 Data and Header Bit Error Rate Performance__________________ 69 6.5 Conclusion 71 Chapter 7 72 Bias-Induced Diversity-Detection (BIDD) Technique for Robust Transmission of Subcarrier-Multiplexed Channels_______________________ 72 7.1 Introduction________________________________________________ 72 7.1 Concept_____________________ 74 7.2 Experimental Results_______________________________________ 77 7.2.1 Data Rate Performance_______________ 77 7.2.2 Chromatic Dispersion Induced RF Power Fading_____________ 78 7.2.3 Polarization Mode Dispersion induced Power Fading__________ 80 7.2.4 RF Intermodulation Terms______________ 82 7.3 Conclusion 83 Conclusion ________________ 84 Bibliography ____________________________________________________ 86 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. List of Figures Fig. 1.2: (a) Simple point-to-point link, (b) Dynamic reconfigurable network employing switching nodes to route the signal to its destination. ___ Fig. 1.3: Schematic of a typical subcarrier multiplexed lightwave system. Fig. 1.4: Electronic header processing at a switching node requires the entire data stream to be detected and processed at the bit rate.______ 9 Fig. 1.5: Optical network model that uses subcarrier header and/or labels.___10 Fig. 1.6: The received RF power spectrum after transmitting a double sideband subcarrier multiplexed signal through single mode fiber 11 Fig. 1.7: Chromatic dispersion coefficient D versus wavelength of conventional single mode fiber and dispersion shifted fiber. ______ 14 Fig. 2.1: The concept of residual dispersion slope in fiber optic links_______ 19 Fig. 2.2: Dispersion induces the regeneration of the clock frequency component, which increases in power with increasing dispersion. _____________________________________________ 21 Fig. 2.3: (a) The experimental setup for monitoring the residual dispersion slope evolution versus the transmission distance, (b) Details of the recirculating fiber loop, (c) The dispersion slope monitor. (TOM: Tunable optical filter, PD: Photodetector, NBPF: clock frequency narrow band electrical filter)____________ 22 Fig. 2.4: (a) Regenerated RF clock power vs. dispersion, (b) Measured dispersion value evolution for two DWDM channels along the V ll Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Fig. 3.1: Fig. 3.2: Fig. 3.3: Fig. 3.4: Fig. 3.5: Fig. 3.6: Fig. 3.7: Fig. 3.8: Fig. 3.9: dispersion compensated link, (c) Dispersion slope accumulation obtained from Fig. 3(b)._____________________ 24 Conceptual diagram of the phase diversity technique for distance independent RF fading compensation._________________ 29 RF Fading compensation module detail._______ 32 Experimental setup for distant-independent RF fading compensation ________________________________________ 33 Conceptual diagram of a chirped fiber Bragg grating, where different wavelengths reflect at different positions in the grating. 35 Relative time delay versus wavelength curves for (a) Linearly chirped FBG and (b) Nonlinearly chirped FBG.________________ 36 A nonlinearly-chirped FBG induces a time delay for the two optical sidebands relative to the optical carrier of a subcarrier multiplexed signal._______________________________________ 37 Reflected spectrum of the nonlinearly-chirped FBG for different applied stretching voltages to the piezoelectric transducer.________ 39 Time delay versus wavelength curves of the nonlinearly-chirped FBG for different applied stretching voltages to the piezoelectric transducer.___________________________________ 39 Normalized RF power versus distance for uncompensated and compensated transmission for (a) 8 GHz, (b) 10 GHz and (c) 12 GHz subcarrier frequency. The dashed line shows the theoretical case for RF power fading. ______________________ _41 vui Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Fig. 3.10: Required grating control voltage versus subcarrier frequency to achieve RF fading compensation.___________________________ 42 Fig. 3.11: Fig. 3.12: Fig. 4.1: Fig. 4.2: Fig. 4.3: Fig. 4.4: Fig. 4.5: Fig. 4.6: Fig. 4.7: (a) RF power spectrum of the transmitted subcarrier multiplexed amplitude-shift-keyed data signal, (b) Recovered bit stream after envelope detection____________ 43 Measured BER versus received optical power for different transmission distances of 0 km, 27.7 km and 52.4 km (data rate = 155 Mbit/s, subcarrier frequency = 8 GHz). The insert depicts an eye diagram of the recovered signal.__________________ 44 Conceptual diagram of (a) wavelength conversion of subcarrier channels, (b) DFG in a PPLN waveguide using a c(2):c(2) process______________ 48 Experimental setup. ________________________________ 49 Optical spectrum after wavelength conversion in the PPLN waveguide._____________________________________________ 50 Linearity of DFG: wavelength converted signal power vs. signal power, measured after the PPLN waveguide___________________ 51 RF spectra before and after the wavelength conversion.__________ 52 BER curves of subcarrier multiplexed channels for before and after the wavelength conversion. 6 _ _ _________ 53 Received optical power sensitivities for 10-9 BER vs. wavelength spacing between the input and output data signal._____ 53 ix Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Fig. 5.1: Fig. 5.2: Fig. 5.3: Fig. 5.4: Fig. 6.1: Fig. 6.2: Fig. 6.3: First-order PMD induces a differential group delay in an optical sideband of a SCM signal, which leads to a phase difference in the corresponding received subcarrier signals in the photodetector, possibly causing serious power fading.___________ 55 Experimental setup. (TL: tunable laser, 900: phase shifter, PC: polarization controller, OF: optical filter, Rx: receiver)__________ 57 PMD induced power fading vs. DGD curves for double sideband (DSB-SCM) and single sideband (SSB-SCM) intensity modulation of a 7 GHz subcarrier with and without dynamic first-order PMD compensation, (a) Measurement of 350 independent polarization samples, and (b) Simulation of 10000 independent polarization samples for each modulation format. The solid line corresponds to the theoretical fading penalty for equal polarization coupling into the PSP for first-order PMD (i.e., DGD)._____________________________________________ 59 Measured bit error rate vs. received optical power for 155 Mbit/s DSB-SCM-BPSK and SSB-SCM-BPSK signals at 7 GHz with and without first-order compensation. The measured DGD for both modulation formats is ~40 ps. The inserts show an error-free recovered eye diagram after first-order compensation.___________________________________________ 61 Conceptual diagram for shadow subcarrier multiplexed header transmission and detection for header based optical switching. 64 Experimental setup for shadow subcarrier multiplexed header transmission and detection for switching information.___________ 65 (a-c) Data receiver RF spectrum plots for the shadow SCM only, data channel only, and combined cases, (d-f) Shadow SCM Recovery Module RF spectrum plots for the shadow SCM only, data channel only, and combined cases.__________________ 68 x Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Fig. 6.4: Bit-error-rate curve plots for the effect of the shadow SCM- header channel on the 9.85 Gb/s data channel, (a) Plots for the cases when the SCM-header channel is (i) not modulated, (ii) PSK modulated (iii) ASK modulated, (b) Plots for the cases when the SCM-header frequency is off from the set NL-FBG dispersion value RF fading frequency (7.7 GHz), (c) (i) recovered SCM header, (ii) output of the threshold detector, (iii) 10 Gb/s data stream before the switch, (iv) output of the switch port 0, (v) output of the switch port 1.________________________ 70 Fig. 7.1: Bias-Induced Diversity Detection (BIDD) Concept: Generation of first and second harmonics in the RF domain by beat terms of lower and upper SCM sidebands and optical carrier (c) Total RF power of the different bias method._________ 75 Fig. 7.2: (a) The 1st harmonic RF power fades due to CD while the 2n d harmonic survives, (b) The 1st and 2n d harmonics both fade due to PMD. BIDD method results in a robust optical system, (c) Total RF power considering both CD and PMD________________ 76 Fig. 7.3: Sensitivity of quadrature bias, minimum bias, and BIDD technique as a function of bit rate___________________________ 77 Fig. 7.4: RF power fading as a function of fiber transmission distance______ 79 Fig. 7.5: RF power fading histograms for quadrature, minimum, and BIDD bias for 8-GHz subcarrier _______________ 80 Fig. 7.6: RF Power fading distribution curves for different bias techniques and 6, 7 and 8-GHz subcarriers____________________ 81 Fig. 7.7: RF Spectra for quadrature, minimum, and BIDD bias for two subcarriers at 7 and 8-GHz. __________________ 82 xi Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Abstract The overall performance of non-static analog and digital optical communication systems and networks may be degraded for various reasons. For instance, chromatic dispersion of standard single mode fiber causes pulse distortion for digital transmission systems and RF power fading in analog fiber-optic links. Polarization mode dispersion also affects the microwave signals in a manner akin to that of multipath fading in wireless transmission. These effects can result in unacceptable power penalties and even complete loss of signal. As data speeds continue to increase, latency at switching nodes due to O-E-O conversion is becoming a major bottleneck in optical networks. Moreover, it may become impractical and overly expensive to use electronics at each node of a network to detect the data, process the header information, and retransmit at rates of 40 Gb/s and higher. Subcarrier multiplexed, and low bit-rate header or label signals may prove to be the required mean to overcome these obstacles. To combat the mentioned performance degrading effects, all-optical techniques are highly desirable to enable high-speed on-the-fly processing, which would be essential for future high throughput and dynamically reconfigurable optical networks. xii Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. This paper will present the following experimental demonstrations to enhance the system performance of such optical networks: (1) doubling the usable spectral bandwidth and number of channels in subcarrier-modulated data transmission over optical fiber; (2) Dynamic dispersion slope monitoring for accurate and continuous dispersion and dispersion slope compensation; (3) Distance-independent RF fading compensation using a tunable nonlinearly-chirped fiber Bragg grating; (4) Wavelength conversion of subcarrier channels using difference frequency generation in a PPLN waveguide; (5) Statistics of PMD-induced power fading for double sideband and single sideband subcarrier-multiplexed signals; (6) Dispersion Division Multiplexing for In-Band Subcarrier-Header-Based All-Optical Packet Switching; (7) Bias-Induced Diversity-Detection (BIDD) Technique for Robust Transmission of Subcarrier-Multiplexed Channels. Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Chapter 1 Introduction and background Optical fiber communication systems thrive on the very high bandwidth capacity and low-loss characteristics of single mode fiber. Fig. shows the low-loss regions of single mode silica fiber centered around 1.3 pm and 1.55 pm. The available optical information bandwidth in the 1.55 pm transmission window is approximately 25 THz, which is predominantly used in telecommunication systems. Wavelength division multiplexing (WDM) is the prime candidate to take advantage of the immense available bandwidth of optical fiber, by spreading the transmitted information in the wavelength domain [1], [2]. Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. WDM = w avelength-division multiplexing ■ 25 1 c o 3 THz < S 0.4 r o CD _ Q ! § 0 2 o 10 25 THz WDM c h a n n els 5 1.7 1.6 1.5 1.4 1.3 1.2 W a v e le n g th , fjm Fig. 1.1 : Attenuation spectrum of single mode silica fiber. The insert shows a typical gain spectrum for an erbium-doped fiber amplifier (EDFA). WDM technology uses multiple optical carriers at different wavelengths multiplexed onto the same fiber, each modulated with independent high speed data signals. The combined throughput of several Tbit/s has been reported for such a WDM system [3], A second very import milestone was the invention of the erbium-doped fiber amplifier (EDFA) [4], [5], which enables lightwave transmission over trans-oceanic distances [6], Without EDFA's, electronic regeneration of the optical signal is essential to overcome the loss along the transmission path. Electronic regeneration includes optical to electrical conversion of the optical signal, followed by retiming, reshaping and retransmission of the original information. An EDFA consists of a short length of erbium doped fiber coupled to a pump laser operating at a wavelength of 980 nm or 1480 nm. The pump laser excites the erbium ions to a higher, 2 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. metastable energy state. An optical signal, located in the 1.55 pm wavelength region, passing through the erbium doped fiber will induce stimulated emission, which causes the excited ions to fall back to their ground state and to generate a new photon at the same phase and wavelength as the traversing signal. Another advantage of EDFA’s is the capability of amplifying multiple WDM channels simultaneously, independent of the bit rate and data format of each signal. 1.1 Non-static fiber-optic communication links The simplest transmission network architecture is the point-to-point link, where for instance, point A is directly connected via a designated transmission medium with point B and point C (see Fig. 1.1a). In an optical communication system, optical fiber serves as the transmission medium carrying one or, in case of a WDM system, several optical data channels. Such a simple link seems static at first, but aging and environmental variations can alter the characteristics of the employed devices, such as signal lasers, EDFA's and passive devices. Even small changes can result in degradation of the signals and severely limit the overall performance of the link. Passive compensation schemes to combat the link dynamics might not function properly, therefore tunable techniques need to be employed, to warrant the signal integrity and performance of the link. Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Switching/Routing Node (a) (b) Fig. 1.1: (a) Simple point-to-point link, (b) Dynamic reconfigurable network employing switching nodes to route the signal to its destination. A typical point-to-point link may interconnect local, metropolitan or wide area networks (LAN, MAN or WAN), with a user base varying from a few tens to several hundred-thousands. The demand for the available bandwidth may also increase; requiring an expansion in information capacity, by boosting the data rate or assigning additional wavelength channels. This request for more capacity should be met in a dynamic way to allocate more bandwidth only when needed. To better utilize the available bandwidth and installed hardware resources, a more complex network topology is necessary. Such a topology should by dynamic and reconfigurable to satisfy the demands and requirements of future communication systems [7]. Fig. 1.1b depicts a dynamic reconfigurable network, where switching and routing nodes determine the data path between two points (circuit-switched or packet-switched networks). The length and the characteristics of the data path can vary significantly, depending on how the signal is routed to its final destination. Those variations can 4 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. potentially be more extreme than in a point-to-point link, having much larger consequences on the channel performance. For instance, when a signal going from point A to point B switches to another path, the accumulated chromatic dispersion may be totally different because either the transmission distance changed or the new path employs fiber with dissimilar amounts of chromatic dispersion. The techniques to reduce and compensate the channel the degrading effects in such switched networks have to be actively tunable to be able to react to the dynamically changing environment. 1.2 Subcarrier multiplexing in optical communication systems Wavelength division multiplexed optical communication systems utilize intensity modulation of the light source and direct detection at the photodiode (IM-DD) to transmit and receive information. This information can consist of digital and/or analog signals. In a pure digital system, multiple lower speed data streams are time division multiplexed (TDM) and the resulting baseband transmitted at the high data rate (i.e. 10 Gbit/s). A disadvantage of TDM is that each receiver has to operate at the high data rate, even if the user at that receiver is only interested in only a few of the lower speed data streams. This problem is especially evident in a personal communication access network, which serves as a fiber-optic backbone for wireless applications [8]. In such a network the number of users is very large and may dynamically vary over time, and also the data rates are low (<1 Gbit/s) compared to a 5 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. digital system. The technique of subcarrier multiplexing (SCM) can be used to better utilize the available modulation bandwidth [9], [10], [11]. A SCM fiber-optic communication link employs several modulated microwave subcarriers transmitted over single mode fiber. A diagram of a typical SCM lightwave system is shown in Fig. 1.2. The analog or digital data is electrically up- converted onto the microwave subcarrier frequency using a microwave mixer. Several modulated subcarriers are combined together and this composite signal can either intensity modulate a semiconductor laser or externally modulate a cw laser source utilizing a Mach-Zehnder modulator. After transmission through single mode fiber, the signal is directly detected with a photodiode and pre-amplified. If only a single channel needs to be recovered, a tunable oscillator and microwave mixer can simultaneously select the SCM channel and down-convert it to baseband. Note that there is usually at least another conversion step to an intermediate frequency involved, but it is omitted for simplicity. Also, several SCM channels can be demodulated at the same receiver by using multiple down-conversion mixers driven with the appropriate subcarrier frequency. In addition, the up- and down-conversion is not limited to the RF domain. It can also be accomplished with all-optical techniques, by using external modulators as optical mixers. Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Analog or Digital Data < 5 H x ) VCO f Combiner Recovered Signal SMF , .. . Mixer Low Noise Amplifier Photo Detector Laser Fig. 1.2 : Schematic of a typical subcarrier multiplexed lightwave system. Typical applications for fiber-optic SCM systems include cable television (CATV) and video on demand, antenna remoting and the fiber backbone in a wireless network, and multiuser, interactive LANs. Furthermore, SCM signals can be combined with baseband data, if the lowest subcarrier frequency is located beyond the baseband data. It is useful for separating baseband payload data and low-speed control signals with routing and timing information [12], [13]. This enables the independent detection and processing of the SCM control channels from the baseband data. Employing SCM in fiber-optic links has various key advantages, such as utilizing the large available bandwidth of lasers, modulators and detectors, without being limited by the bandwidth of TDM electronic circuitry. Dividing the broad power spectrum into a group of narrowband channels allows the receiver to operate at much lower 7 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. data rate, which increases the receiver sensitivity and lessens the high-speed requirements in the processing electronics [14]. On the other hand SCM has several disadvantages, the most important being the issue of source nonlinearity. This especially poses a problem when many subcarriers are transmitted from a single source. Secondly, in digital systems SCM requires a larger bandwidth per channel than TDM. Furthermore, despite operating at lower channel data rates, SCM systems perform at much higher frequencies in the millimeter and microwave regions. It is basically a tradeoff between the possibility of accessing more of the laser and detector bandwidth with SCM systems and occupying a larger bandwidth portion to transmit the same amount of information. 1.3 System issues in analog and digital fiber-optic links Digital as well as analog signals are been used extensively in fiber-optic transmission links. Certain effects in optical communication systems, such as chromatic dispersion can severely degrade the signal integrity and ability for error free recovery of the transmitted data. Furthermore, network specific problems such as output-port contention in switching and routing nodes can reduce system performance and limit the throughput in dynamically reconfigurable optical networks. The following part will address particular issues in analog and digital optical communication systems 8 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 1.3.1 Subcarrier Based All Optical Switching As data speeds continue to increase, latency at switching nodes due to O-E-O conversion is becoming a major bottleneck in optical networks. Moreover, it may become impractical and overly expensive to use electronics at each node of a network to detect the data, process the header information, and retransmit at rates of 40 Gb/s and higher (Fig. 1.4). Fig. 1.3 : Electronic header processing at a switching node requires the entire data stream to be detected and processed at the bit rate. Electronic processing poses the additional problems of being modulation-format and bit-rate dependent. Therefore, there is great interest in finding a way to allow a flow of data to traverse the network with little or no header processing until it reaches the network edge. Adding a subcarrier to each WDM channel that contains the routing information for that channel is one of the more promising means for enabling low speed header processing without having to detect the high speed data (Fig. 1.5). The subcarrier may contain simple Mb/s data that is easily received and detected from a tapped off data stream at a networking node. The subcarrier information is independent of the bit-rate and data format, so it is easy to upgrade the data formats IllATAlllHljilA T A i^ l Q Fiber delay > Optical t Switch I dataBB DATA H: Header J Electronic > Header Recognition 9 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. used in the network without having to ehange the header processing modules. The data can be RZ, NRZ, solitons, etc. User G roup C ore O ptics o \ ( o x e xc (A » M O ptical .Term inal nd IFsers M etro N etw ork O ptical Access Node O ptical Access Node Detailed S u b carriers used for routing at the n etw ork edge Sim pie Label S u b carriers used for fast routing through the core Fig. 1.4: Optical network model that uses subcarrier header and/or labels. 1.3.2 RF fading An increasing number of applications require transmission of analog or digital subcarrier-multiplexed (SCM) RF channels over fiber. The millimeter- and micro wave frequency bands offer adequate bandwidth for future high-capacity broadband wireless and local area networks (LAN). One application would be the efficient distribution of millimeter-wave signals from a central office to remote antennas via a fiber-optic link. However, fiber induced distortions can ultimately limit the performance of such a SCM lightwave system [12], especially fiber's chromatic dispersion can critically degrade a channels signals integrity [13]. In case of a transmitted double-sideband (DSB) signal, this frequency-dependent fiber dispersion produces a deleterious time delay between the two transmitted sidebands in the 10 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. optical domain, causing serious RF power fading that is a function of subcarrier frequency, fiber distance, and accumulated dispersion [14]. Fig. 1.5 : The received RF power spectrum after transmitting a double-sideband subcarrier multiplexed signal through single mode fiber Fig. 1.6 shows a received RF power spectrum versus the transmitted distance for a DSB-SCM signal. It can be seen, that the RF power vanishes periodically over the transmission distance, due to chromatic dispersion. Subcarrier Fig. 1.5: The received RF power spectrum after transmitting a double-sideband subcarrier multiplexed signal through single mode fiber Unfortunately, many potential applications involve reconfigurable optical paths, where RF fading dynamically changes with transmission distance. For robust RF- based optical systems, distance-independent RF fading compensation is highly desirable. An experimental demonstration for such a distance-independent compensation scheme will be presented in chapter 4 [32]. Single Mode Fiber RF Power Distance 11 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 1.3.3 Nonideal fiber characteristics Silica fiber exhibits several nonideal characteristics, which can limit the performance of optical communication systems. Besides the attenuation spectrum of single mode fiber (shown in Fig. 1.1), chromatic dispersion and the fiber’s nonlinear behavior can cause severe restrictions in lightwave transmission systems. 1.3.4 Chromatic dispersion Fiber’s chromatic dispersion emerged as a major limiting factor in optical communication systems, after the advent of the EDFA successfully neutralized fiber loss. Chromatic dispersion, or group velocity dispersion, is caused by the frequency dependence of the refractive index n(co) of the dielectric waveguide, i.e. silica fiber. This leads to a variation in the group velocity of the spectral components of the optical signal with each frequency traveling at a slightly different speed through the fiber, therefore causing pulse broadening of the lightwave signal. Consequently, this signal distortion can severely impair the system performance. The effects of fiber dispersion can be expressed by expanding the mode-propagation constant ( 3 in a Taylor series around the center frequency w0 [16]: (1.1) fi(<o) = n(a>)~ = /30 +/3l(6>-6>0) + ^-j32(6)-co0)2 + ....... c 2 12 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. where C O = (0Q The pulse envelope travels at the group velocity Vg= l/0i, while the coefficient 02 is responsible for pulse broadening. The wavelength for which 02 = 0 is often referred to as the zero-dispersion wavelength X o . Conventional single mode fiber (SMF) has a zero-dispersion wavelength Xq= 1.3 pm. A more commonly used system parameter with c being the speed of light. For SMF, the dispersion coefficient D is about +17 ps/(nm*km) in the 1.55 pm wavelength region. In general, the dispersion parameters can be changed by tailoring the waveguide profile of the fiber. For instance, such a specialty fiber with a smaller mode field parameter, known as dispersion shifted fiber (DSF), has a zero dispersion wavelength A ,o« 1.5 pm and D varies between -2.5 and +2.5 ps/(nm*km) around A , < > . Fig. 1.7 depicts the dispersion parameter D as a function of wavelength for conventional SMF and DSF. for chromatic dispersion is the group velocity dispersion coefficient D and it can be expressed as: (1.2) dX vg dX dX d a £ 13 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 2 0 Conventional -- 17 Dispersion Shifted W avelength (pm) 1 . 1 1. 2 /1 .3 Sl ope = 0. 08 p s / k m- n m2 S - 10 -20 Fig. 1.6 : Chromatic dispersion coefficient D versus wavelength of conventional single mode fiber and dispersion shifted fiber. The dispersion of optical fiber can vary significantly for different wavelength channels over the EDFA gain bandwidth of ~3 THz This wavelength dependence of chromatic dispersion is known as the dispersion slope (dD/dX) or second-order dispersion. A typical value for the dispersion slope for SMF and DSF is around 0.08 ps/(nm2*km). As mentioned before, dispersion induces pulse broadening in a modulated optical signal, which in turn can cause inter-symbol interference (ISI). This interference imposes a limitation on the maximum transmission distance without the need of regenerating the original optical signal. The maximum dispersion limited distance can be approximated by determining the transmission distance after which a pulse 14 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. has been broadened by one bit time. For an intensity modulated signal, carrying non return to zero (NRZ) data, the estimated dispersion limited distance Ld is given by [17]: ft T 1 c ^ L° ~ BDAZ = B 2DA2 where B is the data rate, AX is the spectral width and X is the wavelength of the optical signal. The second term for Ld in equation (1.3) is valid for single mode lasers with a relatively large spectral width AX compared to the modulation data rate B. The dispersion limited distance Ld is approximately 70 km for transmitting 10 Gbit/s NRZ data over SMF at an optical carrier wavelength of 1.55 pm, therefore requiring a dispersion compensation scheme for long distance optical systems. Such a compensation scheme should be tunable [18], to effectively correct for different amounts of accumulated dispersion occurring in dynamically reconfigurable optical networks. 1.3.5 Nonlinearities The use of dispersion shifted fiber can minimize the consequences of chromatic dispersion on an optical signal along a transmission link, but various nonlinear effects can also accumulate and cause severe limitations. Under conditions of high optical power and long interaction length, the nonlinear behavior of silica fiber can produce degrading effects such as attenuation, distortion and cross channel interference. This can place constraints on the channel spacing, the maximum 15 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. allowed optical power of each channel and may also limit the maximum data rate in WDM systems. To complete the discussion of fiber characteristics, a qualitative overview of nonlinearities in fiber is given. Four basic nonlinearities exist in fiber [19]: (1) Stimulated Raman scattering (SRS), (2) Stimulated Brillouin scattering (SBS), (3) Self- and cross-phase modulation (SPM and XPM) and (4) Four wave mixing (FWM). The first two nonlinearities are caused by stimulated scattering effects within the transmission medium, which manifests itself in intensity dependent gain or loss. SRS transfers a small fraction of optical power from one channel at a shorter wavelength to longer wavelength channels. The channels interact with each other through a vibrational wave as they propagate in the forward direction through the fiber. The shorter wavelength channels experience loss, while the longer wavelength channels gain power. SRS can impose limitations on the maximum allowed number of channels in long-haul systems. SBS can also cause a frequency shift of an optical signal, due to interaction with an acoustical wave. A backwards traveling optical wave is generated, which causes optical power degradation in the affected channel. SBS is mostly critical for lightwaves with very narrow spectral widths (<20 MHz). The last two nonlinear effects derive in general from the nonlinearity of the index of refraction in silica fiber, which is dependent on the intensity of the optical signal. SPM is caused by fluctuations in the power of an optical signal present in the fiber 16 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. and results in variations of the phase of the same signal. This self modulation can temporally disperse the optical pulse and may lead to spectral broadening. Furthermore, high values of dispersion can exaggerate the nonlinear effect induced by SPM. XPM, on the other hand, is an interaction, via the nonlinear refractive index, between the intensity of one optical signal and the phase of the other signals propagating on different wavelengths in a multiple channel WDM system. XPM can cause asymmetric spectral broadening and, combined with SPM and dispersion, affect the pulse shape in the time domain. FWM is the result of the nonlinear waveguide medium causing two propagating optical waves to beat with each other. This generates new tones as sidebands at the difference frequencies, which can interfere with other channels located at those frequencies. Note, that the FWM products occur only for closely spaced channels located at the zero dispersion wavelength X o of the fiber. 17 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Chapter 2 Inline Dispersion Slope Monitoring of Many WDM Channels Using Dispersion-Induced RF Clock Regeneration 2.1 Introduction The two trends leading to the continued growth in capacity for optical systems are higher channel bit rates and more parallel wavelength-division-multiplexed (WDM) channels. For 310-Gbit/s channel rates in any WDM system for links >150 km, periodic compensation of transmission-fiber-induced chromatic dispersion is required. Such compensation is typically accomplished by using a lumped element that produces negative dispersion, and the most common deployed type is dispersion compensating fiber (DCF). 18 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. £ c Increasing Residual e 0 ) a C O Q jo 3 E 3 O o < Decreasing Residual Dispersion Fig. 2.1 : The concept of residual dispersion slope in fiber optic links Unfortunately, all deployed fiber types produce a wavelength-dependent dispersion value such that each WDM channel will accumulate a different amount of dispersion as in Fig. 2.1. Moreover, the wavelength-dependent slope of DCF's negative dispersion does not inversely match the wavelength-dependent slope of transmission fiber's positive dispersion [36,37]. Add to this scenario the fact that many fiber links are not composed of only one type of transmission fiber end-to-end, and the challenge to know the residual dispersion slope value becomes even greater. In general, the dispersion slope among the WDM channels must be compensated periodically along the link, and this compensation is dependent on data rate, fiber type, and overall system wavelength range [38]. An important constraint in this problem is that the WDM dispersion slope may change over time due to: (i) fiber plant replacement and upgrade, (ii) reconfigurable optical networking [39], (iii) fiber 19 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. aging, and (iv) temperature changes [40], especially for 40-Gbit/s systems. Therefore, dispersion-slope compensation, either in-line or at the receiver, must include some form of dynamic monitoring in order to ensure precise dispersion compensation. 2.2 Concept for Dispersion Slope Monitoring Using Dispersion- Induced RF Clock Regeneration We demonstrate a scheme for dynamic dispersion slope monitoring of many WDM channels [35]. Specifically, we use the power of the RF clock that is regenerated at the detection of an NRZ signal due to the residual chromatic dispersion of the fiber link [41]. The RF regenerated clocks of t wo widely seperated WDM channels are monitored in real time, producing an accurate measurement of the residual WDM dispersion slope. We measured the dispersion slope for a 3.5-nm WDM system covering an 800-km link at intervals of 100 km. The deviation of the experimental results from the actual fiber slope values is <5%, which corresponds to a negligible system power penalty even at 40 Gbit/s. This technique can be easily modified to accommodate RZ signals for which the RF clock degrades with an increase in accumulated dispersion. 20 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Back to Back 20 km SMF 40 km SMF 55 km SMF -38.67dBm 9.96 GHz -50.50dBm 9.96 GHz -40.50dBm 9.96 GHz -76.17dBm 9.96 GHz RF Requency (GHz) Fig. 2.2: Dispersion induces the regeneration of the clock frequency component, which increases in power with increasing dispersion. When an NRZ format data is modulated on an optical channel and transmitted over back-to-back or zero residual dispersion link, the RF spectrum of the detected signal has a negligible clock frequency component. As the residual dispersion of the link increases, we see that the mentioned clock component regenerates, at a positive nonlinear rate dependent on the net dispersion value (Fig. 3.2). From the power of the regenerated clock component, we can determine the residual dispersion for that optical channel. In order to effectively calculate the net or residual dispersion slope that the WDM channels experience, we need two or more dispersion vs. wavelength points. To this end, we utilize the calculation of dispersion value from the regenerated clock component method. We need to monitor only two WDM channels. Our proposed monitoring module dynamically measures the dispersion induced regenerated clock components of two WDM channels, and calculates the dispersion slope value for the WDM system. 21 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 2.3 Experimental Setup Our experimental setup is depicted in Fig. 2.3(a). Two CW laser lights, one at 1555.00 nm and the other at 1558.53 ran, are modulated with 9.85 Gbit/s 215-1 pseudo-random bit sequences. The optical channels are then transmitted through a dispersion-managed recirculating fiber loop. Recirculating | Fiber Loop ! 9.85-Gb/s PRBS Loop Control LD LD MUX EOM Dispersion Slope Monitor & Trigger (a) 2 0 k m SMF Loop Control AOM AOM 84 km EDFA SMF -MkzJVH-g- TOF I NBPF RF Power $,a V e eter Micro Processor Loop Trigger P a i n ' * 6 k m Flattener EDFA DCF EDFA (b) » * Dispersion Slope (c) Fig. 2.3: (a) The experimental setup for monitoring the residual dispersion slope evolution versus the transmission distance, (b) Details of the recirculating fiber loop, (c) The dispersion slope monitor. (TOM: Tunable optical filter, PD: Photodetector, NBPF: clock frequency narrow band electrical filter) 22 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. In order to demonstrate the dynamic dispersion slope monitoring, we utilize a circulating dispersion-managed fiber loop (Fig. 2.3b). The dispersion slope experienced by the WDM channels builds up at every consecutive circulation of the loop, The loop consists of 84 km of SMF, 16 km of DCF, and a series of controlled gain EDFAs to maintain the consistency of the optical power within the loop after each round trip. Acousto-optical modulators (AOM) are used as switches to direct the incoming WDM packets to-and-from the circulating loop. Each loop circulation takes 500 us to complete, and each loop packet has a duty ratio of 4%. The fiber loop has been adjusted to circulate the incoming WDM channels up to 8 cycles, which enables us to observe the effect of dispersion-managed transmission, as long as 800 km, on WDM channels. The details of the dispersion slope monitor are presented in Fig. 2.3(c). The RF power of the regenerated clock component is measured for two WDM channels. Due to the characteristics of the circuits used for the regenerated clock RF power measurement and the photodiode used to monitor the optical power, the measured clock amplitude needs to be normalized to the square root of the optical power. The normalized clock frequency RF power provides us with the accumulated dispersion value of that particular WDM channel. These values are processed in the microprocessor and the residual dispersion slope of the link is calculated. 23 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 2.4 Results The relationship between the normalized clock amplitude and the chromatic dispersion has been investigated for calibrating the dispersion slope monitor module microprocessor (Fig. 2.4(a)). Mathematical interpolation of the measured clock RF power with respect to this data provides us with a measure of the individual WDM channel net dispersion value for each additional loop circulation (Fig. 2.4(b)). In Fig. 2.4(c) we can see the evolution of the residual dispersion slope versus traveled fiber distance. As predicted, the dispersion difference—hence the residual dispersion slope—is positively dependent on the fiber link distance. 520 -45 ■ 1555.00 nm ▼ 1558.54 nm 500 480 : ® -50 460 440 : < o 420 .55 400 380 C > 300 400 500 600 700 800 900 Fiber Transmission Distance (km) Dispersion (ps/nm) Fig. 2.4: (a) Regenerated RF clock power vs. dispersion, (b) Measured — 12 dispersion value evolution for two DWDM channels along the dispersion compensated link, (c) Dispersion slope accumulation 300 400 500 600 700 800 900 Fiber Transmission Distance (km) (C) obtained from Fig. 3(b). 24 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Previously the residual dispersion slope of the loop has been measured to be ~2.3 ps/nm2/loop. From Fig. 2.4(c), the residual dispersion slope for a single loop circulation is calculated as -2.33 ps/nm2/loop, which is rather close to the stand alone measurement. Due to sensitivity limitations, data points for first two circulations are omitted. The resolution of the system (0.5 ps/nm2/100 km) can be improved further by deploying better electrical circuits. This method for monitoring the residual dispersion slope can be used veiy efficiently for controlling dispersion slope compensation subsystems. 2.5 Conclusion We demonstrate an inline dispersion slope monitoring scheme for dynamically measuring the residual dispersion of a dispersion-managed fiber optic link by taking advantage of chromatic dispersion induced RF clock regeneration. From the simultaneous monitoring of two WDM channels’ regenerated clock powers, we deduce the residual dispersions for the channels and the residual dispersion slope of the fiber link. 25 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Chapter 3 Distance-independent RF fading compensation using a tunable nonlinearly-chirped FBG in a phase diversity configuration 3.1 Introduction An increasing number of applications require transmission of analog or digital subcarrier-multiplexed (SCM) RF channels over fiber. However, transmitting traditional double-sideband (DSB) signals is problematic due to chromatic dispersion. This frequency-dependent fiber dispersion produces a deleterious time delay between the two transmitted sidebands in the optical domain, causing serious RF power fading that is a function of subcarrier frequency, fiber distance, and accumulated dispersion [25]. Unfortunately, many potential applications involve reconfigurable optical paths, where RF fading dynamically changes with transmission distance. For robust RF-based optical systems, distance-independent RF fading compensation is highly desirable. Several approaches have been proposed to compensate for RF power fading in conventional DSB systems, including minimum transmission biasing [31], adjustable modulator chirp, and linearly-chirped fiber Bragg gratings [31], but they are all 26 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. dependent on transmission distance and must be actively tuned to accommodate different path lengths. We have previously reported the use of a nonlinearly-chirped fiber Bragg grating to provide tunable compensation for dispersion-induced RF power degradation in variable-length multiple-channel SCM transmission links [30]. This grating has the ability to uniquely provide a tunable dispersion to incoming signals since the time delay as a function of wavelength has a nonlinear shape. However, this technique also suffers from being transmission-length dependent. Single sideband (SSB) transmission, which is immune to the problem of dispersion- induced power fading, has also been proposed [45]. This technique can be complicated to implement, and additionally, in a multiple-channel SCM system each channel must have its own SSB-generating circuitry. We use a nonlinearly-chirped fiber Bragg grating (FBG) in a phase diversity configuration to achieve distance-independent RF power fading compensation for DSB subcarrier-multiplexed systems [32]. In this technique, the incoming signal is split into two components, and a □ phase shift is induced in the sidebands of one arm relative to the other by reflection off the stretched nonlinearly-chirped FBG. At the output of this phase-diversity configuration, orthogonally-polarized components of the signals in the two arms are combined to reduce coherent crosstalk effects in the fiber. We demonstrate distance-independent compensation of RF power fading from 0 to 150 km for 8, 10, and 12 GHz subcarrier frequencies, with received RF power in 27 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. all cases flat to within 1 dB. Error-free transmission is also achieved for 0,27, and 52 km transmission of a 155-Mbit/s SCM/amplitude-shift-keyed channel. 3.2 Concept of RF fading compensation using phase diversity As mentioned before, fiber's chromatic dispersion can cause severe RF power fading as a function of subcarrier frequency, transmission distance and accumulated dispersion. Without compensation, the received subcarrier power is a function of transmission distance, as shown in Fig. 1.6, with periodic power drop-outs: (3-1 )P eljRF ^cos where < p l and < p 2 are the phases of the sidebands relative to the carrier, c is the speed of light, L is the transmission distance, D is the dispersion, fRF is the subcarrier frequency, and fopt is the optical carrier frequency for the transmission wavelength X. The periodic length AL at which the drop-outs occur, can be expressed by: f 9\ +<Pi] = cos2 ncLD ( f J RF 2" = cos2 k LDA2 fx F I 2 J opt j c (0.2) AL = D Z flp The distance Li where the first power drop-out occurs is given by: (3-3) I, = ----- %-r 2 D t f e To give an example, a 10 GHz subcarrier transmitted on an optical carrier at a wavelength of 1.55 pm over standard single mode fiber with a chromatic dispersion of +17 ps/nm*km experiences complete extinction after a transmission distance of 28 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. -36 km. This demonstrates the severity of the RF fading problem, since fiber spans of >30 km are highly desirable in optical communication systems. The concept behind our distance-independent RF fading compensation technique is illustrated in Fig. 3.1. In our compensation module, the nonlinearly-chirped fiber Bragg grating arm induces an additional relative phase shift of □ between the optical sidebands, resulting in a received electrical power that is again a function of distance, with periodic power drop-outs. Compensation Module RF Power Distance SMF RF Power Distance RF Power Distance Constant RF Power Fig. 3.1: Conceptual diagram of the phase diversity technique for distance independent RF fading compensation. However, the different phases in the two arms cause the power drop-outs through one arm to occur at different distances than the drop-outs in the other arm. Since the incident optical signals are orthogonally-polarized, they do not interact in the 29 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. photodetector. After photodetection, these two optical signals result in a pair of currents, one from the arm containing the grating (Ig): (3.4) Ig o c c, cos j cos (2rfR F t + G g) and one from the other arm (I0 ): (3.5) I 0 O C c2 cos -'1 + cos{l7rfR F t + 0o) V 2 where cl and c2 are constants representing the optical powers from the two arms incident on the photodetector, and 0g and 0O are the phase (or path length) differences between the two arms, respectively. The average electrical power of the superposition of the two currents can be calculated by integrating over one period T of the subcarrier frequency: (3-6) 00 T/2 -T /2 T/2 ■ J -T/2 C jC O S — 12+ j c o s ^ / ^ r + 0g) + c2 C O S ^ 2 JC O S ^ ^ + < 9 °) dt where T = ITrf is the period of the subcarrier frequency. By setting the optical powers in the arms equal and introducing an additional optical delay between the two arms o f the m odule such that 0g — 0O = 7i/2 equation (3.6) can be written as: 30 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. (3.7) J sin \ ( \ ~ P — + — sin(2^r^jp/)+cos — + ^ 2 c o s (2 ^ r/^ ) dt Solving the integral in equation (3.7) leads to the total received subcarrier power proportional to: Equation (3.8) shows that the total received RF power of the combined signal is constant, and therefore independent of the transmission distance. Note that without equalizing the optical powers of both arms and more important, without offsetting the optical path length in the two arms to achieve a 7t/2 phase differential of the two received currents, the total combined electrical RF power would fluctuate depending on the transmission distance. 3.3 Experimental setup of the compensation module Details of the implementation of the RF fading compensation module are shown in Fig. 3.2. The subcarrier signals are transmitted through a certain distance of single mode fiber (SMF), ranging from 0 to 150 km, before entering the RF compensation module. (3.8) = constant 31 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. RF Fading Compensation Module Subcarrier Circulator SMF Coupler J Piezoelectric Stretcher Nonlinearly-chirped FiberBragg Grating Fig. 3.2 : RF Fading compensation module detail. The lengths of the upper and lower arms are phase matched, and an additional D/2 phase shift is applied to one arm to warrant constant received RF power. The splitting ratio at the input to the module is adjusted to compensate for different losses in the two arms. The grating provides the required differential phase shift to the optical modulation sidebands in that arm. It is attached to a piezoelectric transducer (PZT), which stretches the grating to allow tuning to a different subcarrier frequency. The signal bounces of the grating and leaves the circulator out of the third port. A polarization controller and a polarization beam splitter allow the signals from the two arms to recombine with minimum coherent crosstalk. The optical path lengths of the two arms are matched at the required grating voltage, and then the lengths are offset by 1/4 of the period of the subcarrier frequency fR F , which 32 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. corresponds to an additional phase shift of D/2. With this adjustment, the received subcarrier power is independent of the transmission distance. Note that the RF fading compensation module is effective at any location between the transmitter and receiver. In addition, the module can compensate for different transmitter chirps, since the effect of chirp manifests itself in the received subcarrier power as a simple distance change. The experimental setup to conduct bit error rate measurements is drawn in Fig. 4.3. A 155 Mbit/s pseudo random bit stream (PRBS) can be multiplexed onto a 8 GHz subcarrier, to generate an electrical double sideband signal (DSB). This subcarrier multiplexed amplitude-shift-keyed (SCM-ASK) signal is transmitted in the optical domain through a distance of SMF, before arriving at the RF fading compensation module at the receiver. 8 GHz Subcarrier 155 Mbit/s PRBS Data Laser SMF EO-MOD RF Fading Compensation Module l I ► Rx ----- > J Envelope Detector Fig. 33: Experimental setup for distant-independent RF fading compensation 33 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. After compensation and optical to electrical conversion, the original data signal is recovered using envelope detection. Note that the subcarrier is added to the electrical DSB signal by shifting the high level of the PRBS data to 0 V, therefore it enables envelope detection at the receiver. 3.3.1 Nonlinearly-chirped fiber Bragg grating 3.3.1.1 Operating principle An optical fiber Bragg (FBG) grating is a periodic perturbation in the refractive index along the fiber which is formed by exposure of the to an intense optical interference pattern [45]. After the first demonstration of a permanent grating in an optical fiber [46], fiber gratings have emerged as an important device in a variety of optical fiber applications, due to their inherent fiber compatibility, low loss, low cost and polarization insensitivity. Applications of FBGs include optical filtering, wavelength division multiplexing and demultiplexing, fiber-optic sensors, wavelength stabilization in lasers, dispersion compensation and gain equalization [26]. The periodic refractive index perturbation in a FBG basically acts as a filter. Small amounts of the incident optical field can be reflected at the index changes and those partial reflected lightwaves can constructively interfere to generate a strong back- reflected wave. For uniform FBGs this strong interaction occurs at specific 34 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. wavelength known as the Bragg wavelength X b and depends on the refractive index of the waveguide and the grating period. On the other hand, a chirped grating, where the grating period is not constant but monotonically increasing (or decreasing), can reflect different wavelengths at different positions in the grating, as shown in Fig. 3.4. The different time delays of the spectral components of the incident signal, where the short wavelength components travel further into the grating before being reflected, result in pulse compression for the reflected signal. t < - Incident A a R eflected Chirped FBG X j X < 2 ^3 ) ) ) ) ) ) External M echanical Stretcher p : j Fig. 3.4: Conceptual diagram of a chirped fiber Bragg grating, where different wavelengths reflect at different positions in the grating. 35 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. The relative time delay versus wavelength curves for a linearly-chirped FBG is plotted in Fig. 4.5a. When the grating is uniformly stretched with an external mechanical stretcher, i.e. a piezoelectric transducer, the induced time delay curves shifts towards longer wavelength. But the slope of the time delay curves at a certain wavelength A,o, which represents dispersion, does not vary by stretching the grating. To achieve a wavelength dependency of the induced dispersion of a chirped FBG, Linearly-chirped Nonlinearly-chirped Relative * Time Delay (ps) Relative i Time Delay (ps) i k stretch ) \ stretch Wavelength (nm) W avelength (nm) (a) (b) Fig. 3.5: Relative time delay versus wavelength curves for (a) Linearly chirped FBG and (b) Nonlinearly chirped FBG. the grating pitch has to follow a nonlinear function. The time delay versus wavelength curves of such a nonlinearly-chirped FBG are depicted in Fig. 35.b. It can clearly be seen, that the dispersion changes for a constant wavelength as as the grating is stretched towards longer wavelengths. 36 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Relative T im e Delay (ps) At At W avelength (nm ) R F Fig. 3.6: A nonlinearly-chirped FBG induces a time delay for the two optical sidebands relative to the optical carrier of a subcarrier multiplexed signal. If the two optical sidebands of a subcarrier multiplexed signal reflect of a nonlinearly-chirped FBG, they experience a different time delay relative to the optical carrier. Fig. 3.6 shows a time delay versus wavelength curve for a nonlinearly-chirped FBG with two sidebands of a SCM signal located at Xo - fR F and X o + fR p, respectively. To achieve a 180° phase difference between the two sidebands, the combined delay A t* has to add as follows: (0.9) A x , = A xx + A t 2 = - i — J RF where An and At2 are the time delays of the sidebands relative to the carrier after reflecting of the grating, and fR F is the subcarrier frequency. The dispersion value to provide the □ phase shift between the optical sidebands can be approximated by: 37 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. (0.10) D « = ------ = — J— A Arp 2 f ftp A Xgp ^fnp A0 with A X rf being the separation of the optical sidebands in the wavelength domain, and Xo the transmission wavelength. This leads to a necessary amount of ~500 ps/nm of dispersion for a 8 GHz subcarrier signal to achieve a 180° phase difference between the two sidebands in the 1.55 pm wavelength region. Using the nonlinearly-chirped FBG is essential for the phase diversity technique, since operating at different points along the grating enables to achieve different dispersion values so that different subcarrier frequencies can be compensated by applying the □ phase shift between the optical sidebands. This would not have been possible by using a linearly-chirped FBG. 3.3.1.2 Measured grating characteristics The grating used in the setup of the RF fading compensation module is a nonlinearly-chirped FBG [48]. The grating is mounted onto a piezoelectric transducer to provide the mechanically stretching ability. The measurements are conducted with the grating connected to the second port of the 3-port circulator. Fig. 3.7 shows the reflected spectrum of the of the nonlinearly-chirped FBG for different applied stretching voltages to the PZT. The bandwidth of the nonlinearly-chirped FBG is -1 dB and the reflected band can be shifted ~0.8 nm towards longer wavelengths, by applying 500 V to the PZT. 38 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. -35 300 V 500 V -40 ov 1 ) ts -60 -65 1553 1550 1551 1552 1554 Wavelength (nm) Fig. 3.7: Reflected spectrum of the nonlinearly-chirped FBG for different applied stretching voltages to the piezoelectric transducer. Fig. 3.8 depicts the shift in the time delay versus wavelength curves for different applied control voltages of 100 V, 300 V and 500 V to the PZT. 13 G < D 1 500 300 y 400 300 500 V 100 V 200 100 0 1551 1552 1553 Wavelength (nm) Fig. 3.8: Time delay versus w avelength curves o f the nonlinearly-chirped FBG for different applied stretching voltages to the piezoelectric transducer. 39 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. The slope of the time delay curve, which represents dispersion, increases from -300 ps/nm to -900 ps/nm for a fixed wavelength as the control voltage is increased. Note that the curve remains its form as it shifts across the wavelength regime. The operation point on the nonlinearly-chirped FBG can be chosen by tuning the PZT voltage to achieve the necessary time delay slope required for the 180° relative phase shift between the sidebands. Virtually, there is no upper boundary on the subcarrier frequency, which can be compensated for, only the lower frequency may be limited by the maximum available amount of dispersion induced by the grating. 3.4 Experimental results Unmodulated RF subcarriers at frequencies of 8 GHz, 10 GHz and 12 GHz are transmitted over a range of distances up to 150 km to illustrate the performance of the RF fading compensation module from Fig. 3.1. The received RF power versus the transmission distance for the uncompensated and the compensated case is shown in Fig. 4.9 together with the theoretically expected RF fading curves for the three subcarrier frequencies. The theoretical curves for the received RF power are calculated using equation (3.1) on page 28 and the appropriate dispersion values for the employed fiber. The periodic RF power drop outs occur at transmission distances of -65 km for 8 GHz, -36 km and -108 km for 10 GHz, and -26 km, -77 km and -129 km for 12 GHz. This shows that even relative short spans of fiber (<50 km) can impose a severe power penalty in subcarrier multiplexed systems. 40 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Using the compensation module, the received RF power is flat to within ~1 dB for the three subcarrier frequencies. The 3 dB power penalty compared to the best case value for uncompensated transmission for the same received optical power is inherent to the phase diversity technique. It can be explained by the equal splitting I O h & T3 -10 '§ -20 C 4 T3 | -30 I I -40 C om pensated 4 M easured Theoretical 8 GHz 0 50 100 D istance (km ) (a) 1 19 1 2 -10 -20 -30 -40 C om pensated 4 M easured Theoretical J 1 0 G H z I -10 50 100 D istance (km) (b) 150 1 '2° 0 4 n -3 0 1 9 . -40 150 C om pensated — , 4 M easured f / heoreticaU 50 100 D istance (km ) (C) 1 50 Fig. 3.9: Normalized RF power versus distance for uncompensated and compensated transmission for (a) 8 GHz, (b) 10 GHz and (c) 12 GHz subcarrier frequency. The dashed line shows the theoretical case for RF power fading, ratio in the compensation module, which yields the two currents in the photodetector from equation (3.4) and (3.5). Even if one of the current is at its minimum the other is at its maximum, because of the 90° phase difference between the two signals. This 41 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. leads to 50 % less received power than the best uncompensated case, but it is constant for any transmission distance. Fig. 3.10 depicts the required control voltage for the PZT to stretch the grating versus the subcarrier frequency to obtain the 180° phase shift between the optical sidebands, thus achieving RF fading compensation. It displays the necessary monotonous tuning characteristics to accomplish reliable compensation. 00 M 0 > 1 o U 500 450 400 350 300 250 200 8 9 10 11 12 13 7 Subcarrier Frequency (GHz) Fig. 3.10: Required grating control voltage versus subcarrier frequency to achieve RF fading compensation. BER measurements were taken with 155 Mbit/s pseudorandom data, which was electrically up-converted to 8 GHz. The SCM-ASK data signal is transmitted and envelope detected at the receiver. Fig. 3.11 a shows the RF power spectrum of the 42 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. SCM-ASK signal before transmission and Fig. 3.11 b displays recovered data bits after envelope detection at the receiver. The BER versus received optical power curves for fiber spans of 0 km, 27.7 km and 52. km are plotted in Fig. 3.12. Error free transmission is achieved for the three transmission distances, where 0 km corresponds to maximum RF power fading in the grating arm of the module, while 52.4 km corresponds to almost maximum RF power fading in the non-grating arm. RF Power Spectrum Recovered Data 7.5 8.5 8 0 10 20 30 40 50 Frequency (GHz) Time (ns) (a) (b) Fig. 3.11: (a) RF power spectrum of the transmitted subcarrier multiplexed amplitude-shift-keyed data signal, (b) Recovered bit stream after envelope detection 43 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 2 3 4 5 6 7 8 9 10 0km 27.7 km 52.4 km Baseline -13 -12 -11 -10 -9 -8 -7 Received Optical Power (dBm) Fig. 3.12: Measured BER versus received optical power for different transmission distances of 0 km, 27.7 km and 52.4 km (data rate = 155 Mbit/s, subcarrier frequency = 8 GHz). The insert depicts an eye diagram of the recovered signal. The BER performance does not depend on whether the received electrical subcarrier power passes through the grating arm or the non-grating arm, therefore resulting in distance independent RF fading compensation. Most of the -2.5 dB optical power penalty relative to the back-to-back BER measurement comes from the 3 dB electrical power penalty inherent in the phase diversity technique. The asymmetric duty cycle of the recovered eye diagram shown in the insert of Fig. 4. 12, was caused by the nonideal characteristics of the employed envelope detector, which was not optimal suited to recover digital amplitude modulated signals. 44 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 3.5 Conclusion We demonstrated the use of a nonlinearly-chirped fiber Bragg grating in a phase diversity configuration to achieve distance-independent RF power fading compensation for double sideband subcarrier multiplexed fiber-optic communication systems. RF power fading compensation for distances ranging from 0 to 150 km and for subcarrier frequencies of 8, 10, and 12 GHz was shown. The received RF power was flat to within ~1 dB in all cases. Furthermore, BER measurements of transmitted 155 Mbit/s PRBS data, subcanier multiplexed to 8 GHz, proved error free recovery of the received signal with an optical power penalty of ~3 dB, which is inherent to our technique. 45 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Chapter 4 Wavelength Conversion of Subcarrier Channels using Difference Frequency Generation in a PPLN Waveguide 4.1 Introduction For many applications, it is quite advantageous to transmit several analog or digital subcarrier-multiplexed (SCM) RF channels over an optical fiber link or network. These applications include: cable TV, wireless network interfaces, microwave photonic systems, and control information for optical networking and optical packet switching [21,53]. Moreover, subcarrier modulation is important for data grooming, bandwidth allocation flexibility, and access networks. Next generation networks may have routing and switching capabilities in the wavelength-division-multiplexing (WDM) layer for flexible bandwidth allocation. Reconfiguration can be achieved either by a tunable transmitter or a tunable receiver. In either case, wavelength conversion is needed in order to increase the efficient use of the limited wavelength pool as well as resolve wavelength contentions [54]. It is only natural to envision a future platform that combines subcarrier multiplexing with wavelength conversion for high-throughput network performance. In such a 46 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. network, wavelength conversion should support signals of arbitrary formats/protocols in order to accommodate various user applications and facilitate network interoperability. All-optical wavelength shifting of subcarrier signals can be accomplished by several methods, including: cross-gain modulation (XGM), cross phase modulation (XPM), four-wave-mixing (FWM), and difference-frequency generation (DFG) [55-58]. However, only FWM and DFG offer complete format transparency. Moreover, all methods except DFG generally use a semiconductor optical amplifier (SOA) as their wavelength-shifting medium, which generates at least one of the following disadvantages, depending on the specific method used: (i) distortions due to nonlinear and population-dependent memory characteristics, (ii) crosstalk, (iii) limited conversion speed, (iv) limited wavelength range, (v) limited conversion efficiency, (vi) additive noise, (vii) spectral distortion, and (viii) limited dynamic range. Alternatively, DFG achieves memoryless, linear, transparent wavelength shifting with extremely low crosstalk and quantum limited additive noise over a wide wavelength range. Wavelength shifting of subcarrier channels has been demonstrated using DFG in an AlGaAs device [59]. 4.2 Experimental Setup In this paper, we demonstrate and characterize a transparent all-optical wavelength conversion process for subcarrier-multiplexed channels in which a memoryless c(2):c(2) DFG process uses 1550-nm pumping in a periodically poled lithium niobate (PPLN) waveguide [52]. We achieve penalty-free all-optical wavelength conversion 47 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. of two 55-Mbit/s subcarrier channels, and the process shows a >30-dB linear dynamic range for crosstalk-free, transparent operation. We note that the PPLN structure has a lower insertion loss than the AlGaAs device in [59]. Fig. 4.1: Conceptual diagram of (a) wavelength conversion of subcarrier channels, (b) DFG in a PPLN waveguide using a c(2):c(2) process Figure 4.1a illustrates the desired wavelength conversion function. Data is imposed on modulated sidebands around an original carrier wavelength 11. After wavelength conversion, these same sidebands are located around a new carrier wavelength 12. Figure lb shows a conceptual diagram of the operation of the c(2):c(2) process used to perform this function [62]. The first c(2) process involves the CW pump (lpump) (a) A ll-O ptical W a v elen gth Converter V. (b) 48 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. undergoing second harmonic generation (SHG) to produce a local oscillator at lpump/2. This then mixes with the modulated signal lsignal(t) through DFG to form a wavelength-shifted copy of the signal at a wavelength of ~(21pump - lsignal(t)). Both parametric processes are instantaneous, permitting modulation bandwidths in excess of several THz. The conversion efficiency is symmetric in the forward and backward directions. Since the DFG conversion efficiency is not proportional to the signal power, the process is linear over a large dynamic range. Moreover, there is no crosstalk sideband at (21signal(t)-lpump) as there is in FWM. EDFA ^-signal EOM H > WDM PPLN ■ K m u BPSK Modulator BPSK Modulator ( |) RF, Q d> RFi ^ D u n r o EDFA BP filter 4 - BPSK @ Demodulator *• 2 ^ p u m p " ^ sig n a l t Data RF| or RF2 Out Pattern Generator Fig. 4.2: Experimental setup. Our experimental setup is shown in Fig. 4.2. A tunable laser is set at 1555-nm and used as a CW signal. A 55-Mb/s on-off-keyed bit stream is binary-phase-shift-key (BPSK) modulated onto an RF1 carrier at 650 MHz, and a second 55-Mb/s channel is modulated onto an RF2 carrier at 1050 MHz. The subcarrier channels are fed into a 2-GHz-bandwidth LiNb03 external modulator. The EDFA is placed after the external modulator and is used to control the signal power launched into the PPLN 49 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. waveguide. A tunable narrow-linewidth extemal-cavity laser at 1550 nm is used as the pump source for initiating the DFG process. The 1550-nm pump source is amplified to +22 dBm and fed into a wavelength-selective coupler. This coupler combines the pump and signal and also filters out the amplified-spontaneous- emission noise of the high power EDFA. At the output of the PPLN waveguide, the wavelength-converted signal at 1545 nm is optically filtered and received. The receiver is connected to a BPSK demodulator to recover the original bit stream. The BPSK demodulator is driven by either a 650-MHz or a 1050-MHz RF carrier to select between the subcarrier channels. 4.3 Experimental Results W avelength converted Signal h i 1545 mil 1550 nm 1555 nm Wavelength (nm) Fig. 4.3: Optical spectrum after wavelength conversion in the PPLN waveguide. 50 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. The optical spectrum at the PPLN waveguide output is shown in Fig. 4.3. The power difference between the 1555-nm original signal and the 1545-nm wavelength-shifted signal at the output of the PPLN waveguide is the conversion efficiency. For a 1550- nm input pump power of ~ 100 mW, a conversion efficiency of -21 dB is observed. One reason for the low conversion efficiency in this particular device is anomalous loss in the fiber pigtails. Typical fiber-to-fiber insertion loss is 3.5 dB [60], while the device used in this work showed a much higher loss of ~ 7 dB. It should also be noted that recent demonstration of a novel buried waveguide design and fabrication technique have led to a threefold increase in the internal conversion efficiency [61]. An optimized device using this technique would allow for 0 dB wavelength conversion with only 75 mW pump power. t; S 1 8 S 3 w o g J3 ? & i£ s 5 1.1 5 10 15 -20 -25 -30 -35 -40 15 -10 -5 0 5 10 15 20 Signal Power (dBm) Fig. 4.4: Linearity of DFG: wavelength converted signal power vs. signal power, measured after the PPLN waveguide 51 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Figure 4.4 shows the variation of the wavelength-shifted output signal power as a function of the input signal power, for which the pump power is kept constant. Wavelength-conversion efficiency of -21 dB is maintained regardless of the input signal power level, demonstrating the large linear dynamic range of the DFG process as expected from theory. To further characterize the linearity of the DFG process, we have measured the RF spectra of the subcarrier channels before and after the wavelength-conversion process. a B efore X conversion S3 vS - U t % < 2 H I A fter X conversion '^ v w y A v y / f l f V RFi RF 2 RF F requency Fig. 4.5: RF spectra before and after the wavelength conversion. As shown in Fig. 4.5, the spectra are virtually identical, indicating transparent wavelength conversion. In contrast, a nonlinear process would show harmonics and spectral distortion. 52 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. P 4 U J P Q '■ w ' b O O ORFi Before X shift A R F 2 Before X shift RFi After X shift RF2 After X shift -34 -33 -32 -31 -30 Optical Power (dBm) Fig. 4.6: BER curves of subcarrier multiplexed channels for before and after the wavelength conversion. 6 Figure 4.6 shows the bit-error-rate (BER) curves of the data signals before and after wavelength conversion for both subcarrier channels. No significant power penalty is observed. To measure the effect of the wavelength spacing between the input signal and the pump, the signal wavelength is varied across a broad range. ^ - 2 1 - -28 £>-29 I -30 C O S -31 co O RFi Before X conversion □ RF2 Before X conversion • RFi After X conversion ■ RF2 After X conversion -20 -10 0 10 20 Wavelength Conversion Distance (nm) Fig. 4.7: Received optical power sensitivities for 10-9 BER vs. wavelength spacing between the input and output data signal. 53 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. In Fig. 4.7, receiver sensitivities for 10-9 BER in terms of optical power are plotted as a function of the wavelength spacing between the input and output signals (i.e., twice the wavelength difference between the input signal and the pump). For both up- and down-conversion, we observe up to 20-nm wavelength shifts with similar performance, limited only by the EDFA bandwidth and filtering characteristics of our equipment. 4.4 Conclusion We demonstrate and characterize a transparent all-optical wavelength conversion process for subcarrier-multiplexed channels. Our memoryless c(2):c(2) difference- ffequency-generation process uses 1550-nm pumping in a periodically poled lithium niobate (PPLN) waveguide. We achieve penalty-free all-optical wavelength conversion of two 55-Mbit/s subcarrier channels. The process shows a >30-dB linear dynamic range for crosstalk-free, transparent operation. 54 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Chapter 5 Statistics of PMD-induced power fading for double sideband and single sideband subcarrier-multiplexed signals 5.1 Introduction Polarization mode dispersion (PMD), due to the random birefringence of single mode optical fiber, is one of the critical challenges in next-generation optical communication systems. A key feature of PMD is its statistical behavior, since the relative orientation between the state-of-polarization (SOP) of the input signal and the principal-states-of-polarization (PSPs) of the fiber varies randomly with time. Slow Polarization i i A Phase Subcarrier Transmitter Square Law Detector Received Subcarrier Power DGD Fast Polarization Fig. 5.1: First-order PMD induces a differential group delay in an optical sideband of a SCM signal, which leads to a phase difference in the corresponding photo-received subcarrier signals, possibly causing serious power fading. Moreover, the differential group delay (DGD) between the fast and slow PSP, i.e. first-order PMD, is a random process with a Maxwellian probability distribution. 55 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. These characteristics of PMD induce a stochastic and dynamically-changing degradation of high speed digital baseband channels (310 Gbit/s). Transmission of analog and digital subcarrier-multiplexed (SCM) signals over fiber will also be severely affected by PMD [63], although its statistical impact on SCM signals has not yet been investigated. Therefore, it is imperative to examine the fading characteristics for SCM signals using a realistic PMD source that closely approximates the statistical nature of PMD. The DGD between the fast and slow PSP of an optical sideband in a SCM signal causes a phase difference in the corresponding received subcarrier signals in the photodetector, as shown in Fig. 5.1. Superposition of the photo-currents may lead to serious power fading of the recovered subcarrier signal due to destructive interference that is a function of subcarrier frequency, accumulated DGD, and optical power splitting ratio between the PSP’s [64]. Furthermore, higher-order PMD can cause additional distortion and degradation of the transmitted signal [65,66]. 5.2 Experimental Setup PMD-induced power fading is similar to the fading that occurs in conventional DSB systems due to chromatic dispersion. Although it has been shown that single sideband (SSB) transmission is relatively immune to chromatic dispersion, it is unclear whether SSB intensity modulation is also beneficial in reducing PMD- induced fading. 56 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Attenuator r iT "' -a < Analyzer > 15-section PMD Emulator Snbcarrier lst-order PMD Compensator Fig. 5.2: Experimental setup. (TL: tunable laser, 900: phase shifter, PC: polarization controller, OF: optical filter, Rx: receiver) We experimentally investigate the statistics of PMD-induced power fading as a function of DGD for DSB-SCM and SSB-SCM signals using a PMD emulator with an average DGD of -40 ps [62]. We find a similar susceptibility to PMD-induced power fading for both modulation formats in the absence chromatic dispersion. A significant improvement in the worst case power fading penalty (-20 dB) is achieved by using a single section of polarization maintaining (PM) fiber in a dynamic first- order PMD compensator. Furthermore, the results of numerical Monte Carlo simulations support the measured data. Figure 5.2 shows the experimental setup used to compare DSB-SCM and SSB-SCM modulation formats under high PMD conditions. Besides conventional DSB intensity modulation using an external single electrode Mach-Zehnder (MZ) modulator, SSB modulation is achieved by employing a dual electrode MZ modulator and driving the second input with a 90° phase shifted copy of the subcarrier signal input to the first electrode [68]. The PMD emulator contains 15 sections of polarization-maintaining 57 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. (PM) fiber, with 9 polarization controllers (PCs) distributed between the sections to realize different polarization coupling and therefore closely emulate the Maxwellian distribution of DGD (measured average DGD -40 ps) [68]. The input signal to the PMD emulator can be selected between the modulated signal and the PMD analyzer output, where a tunable laser is used to determine the actual DGD value of the emulator. The compensated and uncompensated received subcarrier power is measured for the same emulator state and same adjusted optical power by selecting the corresponding optical path. The dynamic first-order PMD compensator consists of an electronically controlled PC followed by a 24 m long section of PM fiber (DGD -42 ps). Some of the light is tapped off after the PM fiber and detected to generate a feedback signal by mixing the received subcarrier signal with itself. The PC maximizes the feedback signal, which is proportional to the received RF power, by optimizing the polarization coupling into the PM fiber. Note, that there is minimal chromatic dispersion in the setup. 5.3 Experimental Results We measured the received subcarrier power and the corresponding DGD values for 350 independent polarization samples by randomly changing the polarization coupling inside the PMD emulator for DSB-SCM and SSB-SCM intensity modulation using a 7 GHz subcarrier. Figure 5.3 (a) shows the RF power fading penalty versus DGD for DSB and SSB formats with and without compensation. The 58 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. solid line in each figure plots the theoretical fading penalty for a 7 GHz subcarrier for equal polarization coupling into the PSP for first-order PMD (i.e., DGD) only. DSB-SCM and SSB-SCM modulation exhibit a similar sensitivity to PMD-induced power fading. Higher-order PMD can lead to a significant fading penalty (>20 dB), as indicated as by some points outside the theoretical fading curve. Unlike for the No Compensation lst-order Compensation 5 30 DSB SSB DSB SSB 0 20 40 60 80 100 0 20 40 60 80 100 0 20 40 60 80 100 0 20 40 60 80 100 120 DGD(ps) DGD(ps) DGD(ps) DGD(ps) (a) No Compensation lst-order Compensation o 30 DSB SSB DSB SSB 0 20 40 60 80 100 0 20 40 60 80 100 0 20 40 60 80 100 0 20 40 60 80 100 1 2 C DGD (ps) DGD (ps) DGD (ps) DGD (ps) (b) Fig. 5.3: PMD induced power fading vs. DGD curves for double sideband (DSB- SCM) and single sideband (SSB-SCM) intensity modulation of a 7 GHz subcarrier with and without dynamic first-order PMD compensation, (a) Measurement of 350 independent polarization samples, and (b) Simulation of 10000 independent polarization samples for each modulation format. The solid line corresponds to the theoretical fading penalty for equal polarization coupling into the PSP for first- order PMD (i.e., DGD). 59 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. case of chromatic dispersion, SSB modulation does not avoid a fading penalty since the PMD effects apply directly to the optical sideband, causing power fading when a single optical sideband is present. The worst case fading penalty can be reduced by -20 dB for both formats by using dynamic first-order compensation. Monte Carlo simulations using Maxwellian PMD statistics were performed to support our experimental data. Figure 6.3(b) shows the simulation results for RF power fading versus DGD for 10000 independent samples. The simulations exhibit a qualitatively comparable performance to the measured data for both formats with and without compensation. The larger maximum fading penalty compared to the measurements may come from the limited number of samples in the experiment. 1% of the samples (100 worst cases) exhibit >14 dB of fading penalty for both formats without compensation. After compensation the 1% power variance is ~4 dB, respectively. Figure 5.4 shows the measured bit error rate (BER) versus the received optical power for a 155 Mbit/s DSB-SCM and SSB-SCM binaiy-phase-shift-keyed (BPSK) signal modulated onto a 7 GHz subcarrier. The baseline is measured without the PMD emulator. The DGD value of the PMD emulator is set to ~40 ps to measure both modulation formats with and without compensation. The power penalty without compensation is >1.5 dB for both formats, compared to <0.5 dB with first-order compensation. 60 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Baseline Baseline Received Optical Power (dBm) Received Optical Power (dBm) Fig. 5.4 : Measured bit error rate vs. received optical power for 155 Mbit/s DSB- SCM-BPSK and SSB-SCM-BPSK signals at 7 GHz with and without first-order compensation. The measured DGD for both modulation formats is ~40 ps. The inserts show an error-free recovered eye diagram after first-order compensation. 5.4 Conclusion We experimentally and numerically compare the statistics of power fading for DSB and SSB subcarrier-multiplexed signals under high PMD conditions (average DGD ~40 ps) with and without dynamic first-order PMD compensation. We find that both SSB-SCM and DSB-SCM signals exhibit similar sensitivity to PMD-induced power fading. First-order compensation reduces the worst case fading penalty by ~20 dB 61 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Chapter 6 Dispersion Division Multiplexing for In-Band Subcarrier-Header-Based All-Optical Packet Switching 6.1 Introduction Packet-switched all-optical networks need to process the header to efficiently route packets to the appropriate destination [69]. One of the most important requirements during the address information extraction for routing purpose is that the header should be able to be processed rapidly and on-the-fly. Subcarrier multiplexed (SCM) header transmission has been proposed as a method to overcome this problem [70- 12}. When the header is subcarrier multiplexed it can be processed at a rate much lower than the data packet bit-rate. This eliminates the need for data-rate-fast digital circuits in cross-connects. Yet the major disadvantages in the standard SCM header transmission are (i) the subcarrier frequency is much higher than the data bit rate, hence higher frequency modulators and receivers are required, (ii) the optical domain bandwidth utilization is inefficient, and (iii) the proposed systems are not easily applicable to commercial transmission systems. 62 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. In this paper we propose a method to transmit SCM-header at a subcarrier frequency lower than the data-rate, specifically at 7.7 GHz for 10 Gb/s data-rate, using dispersion division multiplexed subcarrier transmission [20]. It is important to note that this is not possible using standard SCM transmission, since the subcarrier placed at such a low frequency can’t be filtered out efficiently and this would deteriorate the detected data performance greatly. This technique permits efficient optical bandwidth utilization and results in little power penalty for the received data, since the SCM header channel is invisible with respect to the data channel detector. There isn’t any need for modifying the data receiver architecture, such as adding very narrow band optical receivers or low pass electrical filters. Also to our knowledge this is the first demonstration for SCM-header transmission and switching at 10 Gb/s data-rate. 63 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 6.2 Concept The conceptual diagram for in-band frequency SCM header based switching is depicted in Fig. 6.1. The data channels and SCM-header channels are modulated on the same CW laser signal at different modulators. Then chromatic dispersion is applied to the SCM-header modulated. The effect of dispersion on the SCM-header channel is apparent only after the signal is detected at a photodetector. Applying dispersion to a subcarrier channel introduces a time delay, or phase difference, between the upper and lower modulation sidebands. When the upper and lower sidebands have a 180° phase difference, the sidebands D e te c to r .Data Data Switch 10 Gb/s Data RF Spectrum Laser) Switch Control ! Data + Shadow SCM Header 1 Dispersion i (RF Fading ! - I Recovery) I SCM-Header -M b/s at f, GHz Detection & Down Conversion LI I 3( I C I SI U U “ H jR F Fading) A SCM Shadowed SCM Header RF Spectrum # 1 Spectrum Fig. 6.1: Conceptual diagram for shadow subcarrier multiplexed header transmission and detection for header based optical switching, cancel each other completely at the photodetector. This event is specifically called RF fading [28]. This faded, or “shadowed” SCM signal is combined with the data modulated signal. The “shadow” SCM channel is invisible for the data photodetector, and has no effect on the data bit-error-rate performance. 64 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. In order to recover the shadow SCM-header signal we apply the same amount of dispersion that is used to fade it at the transmission side. This dispersion removes the phase difference between the sidebands of the SCM channel, and by doing so it compensates for the RF fading. Now the header information can be down-converted from the subcarrier frequency and can be used to control the switch in the optical cross-connect. 6.3 Experimental Setup Transmitter Receiver — ^ GHz RF Spectrum 10 Gb/s Data Da Data + DDM- SCM Header jn I I ' * I > [ S w itc h rrm Ts,p — ^ — e DDM-SClVb DM-SCMi Header Switch/ e Control SCM Header SCM -Header Mb/s at % GHz Header Detection & Down Conversion MHz Switch Fig. 6.2: Experimental setup for shadow subcarrier multiplexed header transmission and detection for switching information. 65 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Our experimental setup is shown in Fig. 6.2. Continuous-wave (CW) laser power is split using a polarization-beam-splitter (PBS). On one CW signal we modulate a 9.85 Gb/s PRBS (2A 23-1). On the other CW signal, we modulate the subcarrier multiplexed (SCM) 55 Mb/s binary-phase-shift-keyed (BPSK) header information, with a subcarrier frequency of 7.7 GHz. Standard single-drive electro-optic modulators and commercially-available nonlinearly chirped fiber Bragg gratings (NL-FBG) are used to introduce the dispersion required for RF fading. Both NL- FBGs have a center wavelength of 1546.7 nm with 0.55 nm bandwidth, and the dispersion value range is [-600, -1600 ps/nm]. The amount of dispersion necessary to fade the subcarrier frequency for the "shadow" SCM-header is -1080 ps/nm for 7.7 GHz. The dispersion is applied by reflecting the SCM modulated laser signal from the tuned NL-FBG. The 9.85 Gb/s modulated and the “shadow” SCM-header signals are then combined using a polarization-beam-combiner (PBC). We use polarization beam combination in order to prevent coherent crosstalk between the two signals. The channels are adjusted to have the same optical power after the combination. The combination signal is transmitted over <1 km fiber and inputted to the lithium- niobate switch. Before the switch some portion of the signal is tapped off for the SCM-header recovery module. The recovery NL-FBG in the SCM-header recovery module is tuned to the same dispersion value as the “shadowing” NL-FBG. This compensates for the RF fading of the shadow SCM-header channel. The photodetected signal contains both the data channel and the recovered SCM-header channel. Coherent demodulation is used to obtain the header information. The header 66 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. information is then sent to a threshold detector. The output of the threshold detector is directly used to drive the 1X2 optical switch. This control signal is used to route the incoming signals to the appropriate output ports. The switch output ports 0 and 1 are detected to demonstrate the switching. 67 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 6.4 Experimental Results 6.4.1 Receiver RF Spectra -40 -45 « -50 O C L U. -55 o c -601 : m i i /fait A W -20 -25 I f -30 00 ■ 3 - 3 8 | -40 O -45 a. a. -so o f -55 -60 4 5 6 7 8 9 RF frequency (GHz) (a) 10 4 5 6 7 8 9 RF frequency (GHz) (d) 10 -20 -1 0 -25 -1 5 "E -20 ( D 2. -2 5 S -30 O -35 2. ' 3 5 < 5 -40 § -45 0 . II. -50 0 £ -55 -60 L L -40 -45 -50 10 RF frequency (GHz) RF frequency (GHz) -20 -25 "g -30 ID 2. ‘3 S m -40 O -45 -1 0 -1 5 E" -20 00 2 -2 5 O -35 Q . U. -40 0 0 -45 -50 IL -50 -55 -60 10 10 RF frequency (GHz) RF frequency (GHz) Fig. 6.3: (a-c) Data receiver RF spectrum plots for the shadow SCM only, data channel only, and combined cases, (d-f) Shadow SCM Recovery Module RF spectrum plots for the shadow SCM only, data channel only, and combined cases. 68 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. The electrical power spectra of the data receiver side and shadow SCM-header recovery module are plotted in Fig. 6.3 (a-f). Fig. 6.3(a-c) show the data receiver RF spectrum plots for the cases when (a) only the shadow SCM-header channel is on, (b) only the data channel is on, and (c) both channels are on. Note that the shadow SCM channel is not visible on the power spectrum (shadowed) when both channels are on. Whereas Fig. 6.3(d-f) show the shadow SCM-header recovery module RF spectrum plots for the same cases, respectively. In Fig. 6.3(d) it is seen that the SCM-header channel is regenerated by more than 20 dB, when compared to Fig. 6.3(a). In Fig. 6.3(f) the SCM channel is visible and -17 dB above the noise floor induced by the data channel. 6.4.2 Data and Header Bit Error Rate Performance In Fig. 6.4(b) the effect of using a subcarrier frequency different from the fading frequency is examined. The modulated SCM channel will be less faded as the difference between the SCM frequency and the RF fading frequency for the used NL-FBG dispersion value increases, so as expected the subcarrier frequency change increases the power penalty. The offset in the subcarrier frequencies can also be visualized as offset in the NL-FBG dispersion value. For example, the dispersion difference between the dispersion values required for RF fading 7 GHz and 7.7 GHz is -200 ps/nm, so we can assume that a residual dispersion of -200 ps/nm would introduce another 0.5 dB power penalty. 69 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. a > < * • > as O' u. § U l * > in 3 4 5 as O H • 7.7 GHz ■ 7.0 GHZ ♦ 6.5 GHZ ▲ 6.0 GHZ ■ No Modulation ♦ PSK SCM • ASK SCM 00 8 9 10 -12 -11.5 -11 -10.5 -10 -9.5 12 -11.5 -11 -10.5 -10 -9.5 -9 Optical Power (dBm) (a) Optical Power (dBm) (b) (0 (ii) (iii) m i | iii (iv) (V) Fig. 6.4: Bit-error-rate curve plots for the effect of the shadow SCM-header channel on the 9.85 Gb/s data channel, (a) Plots for the cases when the SCM- header channel is (i) not modulated, (ii) PSK modulated (iii) ASK modulated, (b) Plots for the cases when the SCM-header frequency is off from the set NL-FBG dispersion value RF fading frequency (7.7 GHz), (c) (i) recovered SCM header, (ii) output of the threshold detector, (iii) 10 Gb/s data stream before the switch, (iv) output of the switch port 0, (v) output of the switch port 1. 70 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Time domain waveforms are displayed in Fig. 6.4. (c) for the SCM-header based switching operation, (i) and (ii) are the recovered header and the corresponding threshold detector output, (iii) is a plot of the data stream before the switch, (iv) and (v) are the outputs of the switch ports 0 and 1. It is seen that the switch directs the data stream to the port 1 on for logic “1” output of the threshold detector and to the port 0 for the logic “0” output. 6.5 Conclusion We experimentally demonstrated the viable placement of a 55 Mb/s bit rate header on a 7.7 GHz subcarrier for in band use with lOGb/s data packets [20]. The resultant system is demonstrated to be spectrally efficient. This allowed easy recovery of the header information without the need for faster than data rate photodetectors, and it has the advantage of header being processed with simple, low-speed electronics. We demonstrated that when such a subcarrier header channel is Dispersion Division Multiplexed (DDM) with a 10 Gb/s data channel it has negligible effect on the data channel performance. And at a switching node demonstration, we recover and downconvert the DDM subcarrier header to obtain the routing information. This experiment demonstrates the possible advantage of subcarriers for inline header processing of very high data rate traffic channels. 71 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Chapter 7 Bias-Induced Diversity-Detection (BIDD) Technique for Robust Transmission of Subcarrier-Multiplexed Channels 7.1 Introduction Data transmission using subcarrier modulation (SCM) has many applications, including: CATV, microwave photonics, wireless remoting, and control signaling in optical networks. In general, SCM channels tend to suffer from RF fading, in which the power of the signal fades as a function of the subcarrier frequency and the transmission distance. Such fading can be caused by fiber chromatic dispersion (CD) in the case of double sideband (DSB) transmission, for which each sideband travels at a different speed down the fiber and will periodically become out-of-phase relative to the carrier itself [28]. Moreover, fading can also occur due to polarization- mode-dispersion (PMD) in both single and double sideband transmission[75]. PMD occurs when the speed of light is different between the two polarization axes, and the light from the two axes will periodically be out-of-phase and cancel each other causing fading. In general, it would be highly desirable to be able to transmit SCM channels that do not experience RF fading and that are robust to chromatic dispersion and PMD. 72 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Previous reports of minimizing CD-induced RF fading have included: (i) dispersion compensating fiber Bragg gratings [34], (ii) optical single sideband (SSB) modulation[77], (iii) optical single sideband detection[80], (iv) and minimum bias operation of the modulator [4]. In minimum bias the detected RF signal is at the second harmonic frequency which is robust to CD. However this method has not been demonstrated for SCM data transmission until now. Several schemes that require active polarization tracking and PMD compensation at the receiver end have been studied[81], however PMD-robust transmission has yet to be demonstrated. We demonstrate a technique that induces SCM data diversity during the detection process without using redundant subcarriers or optical channels, or any extra optical bandwidth. By adjusting the DC bias of the modulator, we suppress the optical carrier to a certain level that generates first and second harmonics of the SCM signal at equal RF powers at the photodetecter. In our technique, we employ the total RF power of both the photodetected first and second harmonic SCM signals, thus achieve diversity detection of the SCM data even though we have modulated and transmitted only a single subcarrier signal. It should be noted that we achieve diversity at the photodetector, without any loss, and the system has 3dB better sensitivity than quadrature bias since the optical power is transferred to the subcarriers. We show that our technique is robust to chromatic and polarization mode dispersion. Under worst case CD for a given RF signal, our technique is robust within 6 dB. And against the PMD our technique displays ~8 dB RF power 73 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. improvement for the 5% probability tail compared to quadrature bias for different subcarrier frequencies. 7.1 Concept Fig.7.1 shows the concept of Bias-Induced Diversity Detection (BIDD). After optical modulation, the subcarrier is copied to two sidebands around the optical carrier. For quadrature bias operation, the optical carrier is stronger than the two sidebands, making it the dominant term during the photodetection process thus we only detect the first RF harmonic of the subcarrier. In the case of minimum transmission bias, in which the carrier is suppressed, upper and lower subcarrier sidebands become the dominant terms, and at the receiver they yield a single beating term at the second RF harmonic frequency. For the BIDD bias operation, the optical carrier is reduced to a level that generates first and second RF harmonics of the subcarrier signal with equal RF powers at the photodetector. The second RF harmonic of the subcanier signal is completely robust to dispersion since it is generated by the single beating term of the optical domain upper and lower subcarrier sidebands. The first harmonic of the subcarrier signal is the summation of the beating terms between optical domain upper and lower subcarrier sidebands with the optical carrier. Fig.7.2(a) shows that by adding these two subcarriers we can achieve a dispersion-robust SCM system with at most 3 dB power fading. Fig.7.2(b) shows the RF fading due to PMD for the first and second RF harmonics, which are 74 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. the terms obtained for quadrature and minimum transmission bias, respectively. We will show that by using first and second RF harmonic power, BIDD fades at most 6 dB due to PMD. Fig 7.2. (c) shows the calculated close form formulae for the total RF power for different bias techniques where pis the modulation depth. Modulator Transfer Function lias PMD & CD PD SCM * Modulator Laser (Xc) >own Convert (fs,2fs) DC Bias Quadrature Bias Minimum Bias BIDD Bias Optical Spectra RF Spectra L A fs t Q — 21s fs 2fs Fig. 7.1: Bias-Induced Diversity Detection (BIDD) Concept: Generation of first and second harmonics in the RF domain by beat terms of lower and upper SCM sidebands and optical carrier (c) Total RF power of the different bias method. 75 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. (a) t 1st Harmonic, f RP Fading due to Chromatic Dispersion 2B ( I Harmonic, 2fs Y X r Dispersion (ps/nm) + Quadrature Bias Dispersion (ps/nn^ Minimum Bias 3 dB Dispersion (ps/nn^ BIDD Bias RF Facing due to 1st order PMD (worst case) (b) I f t . Jkl" Harmonic, f, ^ Harmonic, 2fs A **. .♦*’ \ S \ f \ f \ f * * a * * • * * « • * • » » • • \l \l M \l — I ----------- 1--------► + 1 1 1 1 = ---------------:— b . BIDD Total RF Power (\Aa A /\^/\A 6 dB DGD (ps) Quadrature Bias DGD (ps) Minimum Bias DGD (ps) BIDD Bias j ________Total RFJPower____________ 0.5Jo(ax)2 Ji(ax)2 cos2 (xDLCf,f2/fc2) Cos2 (^fDGD) +0.25J,(a;r)4Cos2 (2^-fDGD) BIDD (c) Minimum Quadrature .25Ji(a2T)4 C os2 (2;rfDG D) Jo(«;r)2 Ji(a?r)2 cos(^rZ)ZCf,f2/fc2) 2 C os2 (;rfDG D) Fig. 7.2: (a) The 1s t harmonic RF power fades due to CD while the 2n d harmonic survives, (b) The 1st and 2n d harmonics both fade due to PMD. BIDD method results in a robust optical system, (c) Total RF power considering both CD and PMD 76 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 7.2 Experimental Results 7.2.1 Data Rate Performance In order to compare the BIDD technique with conventional SCM transmission in terms of modulation bandwidth we modulate an 8-GHz RF tone with OOK 21 5 -1 PRBS data, at data bit rates varying from 55 Mb/s to 2.5 Gb/s (Fig.7.3). Due to reduced bias value hence optical carrier power, minimum transmission and BIDD bias operations have 3-dB better sensitivity than the quadrature bias case throughout the different data bit rates. Our BIDD technique displays no distortion or limitation due to the increasing bit rate. -18 Quadrature Minimum BIDD -20 ffl-22 -o -28 -30 0 500 1000 1500 2000 2500 Bit Rate (Mb/s) Fig. 7.3: Sensitivity of quadrature bias, minimum bias, and BIDD technique as a function of bit rate 77 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 7.2.2 Chromatic Dispersion Induced RF Power Fading In order to demonstrate the effect of chromatic dispersion on the SCM system performance, we transmit a 55 Mb/s PRBS 21 0 -1 data SCM signal over 40 to 120 km of single mode fiber (SMF) with an average chromatic dispersion value of 16.5 ps/nm/km. We use three different subcarrier frequencies of 6, 7, and 8-GHz and measure the photodetector power sensitivity at a bit error rate (BER) of lxlO'1 0 for different bias operations of quadrature, minimum transmission and BIDD. Figure-4 shows the RF fading for each subcarrier frequency with respect to the fiber length for different bias operations. The solid line in each figure is the simulation result. Minimum transmission bias case has the best performance since it has a maximum power penalty of 3 dB at different subcarrier frequencies. This is because the RF power at the second harmonic is the result of a single beating term in the case of minimum transmission bias, which in return makes it robust to the dispersion. Maximum RF fading is limited to 6-dB using the BIDD technique. The 3-dB difference between the BIDD and minimum transmission bias is because of the first RF harmonic in the BIDD technique which can fade completely due to chromatic dispersion. When compared to the quadrature bias operation, in which there can be as much as 40 dB RF fading, both minimum and BIDD bias techniques are robust with respect to chromatic dispersion. 78 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. : .M . ■ 1 jl 6 GHz - * - Quadrature ’T i i m i j muin . i.... i.... i.. i»1 1 v 40 60 80 100 120 Fiber Length (km) 7 GHz \ 40 60 80 100 120 Fiber Length (km) 8 GHz - Quadrature Minimum BIDD 100 120 Fiber Length (km) Fig. 7.4: RF power fading as a function of fiber transmission distance 79 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 7.2.3 Polarization Mode Dispersion induced Power Fading We test the PMD effect on the received subcarrier RF powers using a three-section variable-DGD PMD emulator. The emulator consists of three variable DGD elements separated by two fiber-squeezer-based polarization controllers. The DGD values of the sections were changed according to Maxwellian distribution and the polarization rotation between the sections was set to be uniformly random for each PMD data point. The output DGD of this emulator has the same statistical distribution as a real fiber[6]. 8 GHz Subcarrier, Average DGD = 35 ps 350 350 350 Quadrature Bias M inimum Bias 300 300 300 250 250 250 200 200 200 150 150 150 5% Tail 5% Tail 100 100 1 0 0 30 BIDD 5% Tail RF Power Fading Fig. 7.5: RF power fading histograms for quadrature, minimum, and BIDD bias for 8-GHz subcarrier 80 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 6 GHz Subcarrier 7 GHz Subcarrier 8 GHz Subcarrier Minimum ' Minimum — — Quadrature BIDD Minimum — Quadrature BIDD — — Quadrature \ ~ BIDD 99,9 99 95 80 5030 10 1 .1 99.9 99 95 80 5030 10 1 .1 Probability Percent (%) 99.9 99 95 80 5030 10 1 .1 Fig. 7.6: RF Power fading distribution curves for different bias techniques and 6, 7 and 8-GHz subcarriers The average DGD of the PMD emulator is 35-ps, and we use subcarriers of 6, 7, and 8-GHz. For the different bias operations of quadrature, minimum, and BIDD we measure the RF power for 1000 independent polarization samples. Fig. 7.5 shows the RF power fading distribution due to PMD for the different bias techniques for an 8- GHz subcarrier. The average RF-fading for quadrature, minimum, and BIDD bias operations are measured as 9.2, 4.2, and 2.2-dB respectively. Fig. 7.6 shows the RF power fading distribution curves for 6,7, and 8-GHz subcarriers. As it can be seen, the 5% tail for RF fading, which is 26-28 dB in the minimum and 19-22 dB in quadrature bias cases is reduced to 12 dB in the BIDD technique. The 5% tail means that 95% of the samples have lower RF-fading. The results conclude that BIDD bias technique is very robust to PMD effects compared to other bias techniques. 81 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 7.2.4 RF Intermodulation Terms hi -30 43 £ 40 O P i -50 S -60 -70 Quadrature fS j fs- -30 40 " T Z ■ ■ 7 7 - 7 7 + -70 L --— wl Minimum 2fs, 2fs, -30 40 " 5 # -60 7 9 11 13 15 17 Frequency (GHz) , ‘ f ’ 7 9 11 13 15 17 Frequency (GHz) ■► -70 BIDD fSj fs2 2fsj J f A-3 J U - v r H ' q **1 ■ IF1' \ 2fs, fs,+fs2 7 9 11 13 15 17 Frequency (GHz) Fig. 7.7 : RF Spectra for quadrature, minimum, and BIDD bias for two subcarriers at 7 and 8-GHz. Fig.7.7 shows the received RF spectrum for a two-SCM system with subcarrier frequencies at 7 and 8-GHz carrying 55 Mb/s OOK PRBS binary data for different bias techniques with 100 percent modulation depth. In BIDD the inter-modulation terms of 2fsi-fS 2 and 2fS 2-fsi increase by 3 dB when compared to quadrature bias. The minimum transmission and BIDD technique introduce an additional inter-modulation at fsi+fS 2, which is due to beating between pairs of fsi and fS 2 which are at the opposite sides of the optical carrier. It is important to note that this extra term does not affect the subcarrier frequency spacing or data bandwidth since it is located in the middle of the second SCM harmonics which are separated from each other twice the frequency difference of the first SCM harmonics, at 14 and 16-GHz compared to 7 and 8-GHz. 82 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. 7.3 Conclusion We demonstrate SCM transmission and detection that is robust to RF power fading due to chromatic dispersion and PMD [82]. We achieve signal diversity at the photodetector, without any additional optical or electrical channels. We detect both the first and second harmonic at the receiver and reduce the CD fading from 40 to less than 6 dB, and we show ~8-dB improvement for the 5% PMD probability tail. This is also the first demonstration for SCM data detection from the second harmonic subcarriers. 83 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Conclusion In this dissertation, fiber effects such as chromatic and polarization mode dispersions have been investigated and methods for compensation have been demonstrated. A new and novel multiplexing technique, dispersion division multiplexing has been introduced. This technique has the potential to double the number o f subcarrier channels at exactly the same wavelength and subcarrier frequencies. Dispersion induced NRZ data clock tone regeneration has been used in order to facilitate inline and real time dispersion slope monitoring o f a fiber optic link. Also effect o f PPLN based wavelength shifting on subcarrier signals have been investigated. . As proposed in the related proposal, we also investigated the potential o f placing a shadowed DDM header subcarrier within the digital baseband data spectrum. We experimentally demonstrated the viable placement o f a 55 Mb/s bit rate header on a 7.7 GHz subcarrier for in band use with lOGb/s data packets. The resultant system is demonstrated to be spectrally efficient. This allowed easy recovery o f the header information without the need for faster than data rate photodetectors, and it has the advantage o f header being processed with simple, low-speed electronics. We demonstrated that when such a subcarrier header channel is Dispersion Division Multiplexed (DDM) with a 10 Gb/s data channel it has negligible effect on the data channel performance. And at a switching node demonstration, we recover and downconvert the DDM subcarrier header to obtain the routing information. This 84 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. experiment demonstrates the possible advantage of subcarriers for inline header processing of very high data rate traffic channels. In addition to Chromatic Dispersion (CD) Polarization Mode Dispersion (PMD) may have severe implications on millimeter-wave subcarrier multiplexed fiber-optic transmission systems. Those optical systems may serve as fiber backbones for high speed wireless applications, such as wireless local area networks. Mobile and wireless applications are becoming more and more important in today’s dynamic information society. Hence it may be feasible to provide an optical solution to both of these problems at the same time. To this extend, we studied and experimentally demonstrated a novel new technique, Bias Induced Diversity Detection BIDD). By altering the DC modulation bias voltage it is possible to generate an additional second harmonic frequency counterpart to the original subcarrier that is being transmitted. The resultant optical signal has the same optical bandwidth, and we do not need extra electrical signal generators. From the principle of diversity, we also detect and downconvert the second harmonic component. This process proved to robust to CD and PMD respectively within optical 1.5 dB and 3 dB power budgets respectively. 85 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. Bibliography [77] Adamczyk O. H. et al, Microwave Theory and Techniques,Vol.49, pp. 1962- 1967,2001 [62] Adamczyk O.H., Sahin A.B., Yu Q., Lee S., and Willner A.E., “Statistics of PMD-induced power fading for double sideband and single sideband subcarrier- multiplexed signals,” Optical Fiber Communication Conference and Exhibit, 2001. OFC 2001, Volume: 1,2001 Page(s): M05 [16] Agrawal G. 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[20] Sahin A.B., Willner A.E., “Dispersion Division Multiplexing for In-Band Subcarrier-Header-Based All-Optical Packet Switching,” Optical Fiber Communication Conference and Exhibit, 2002. OFC 2002, Page(s): WOl 95 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. [35] Sahin A.B., Yan L.-S., Yu Qian, Hauer M., Pan Z., Willner A. E., “Dynamic Dispersion Slope Monitoring of Many WDM Channels Using Dispersion-Induced RF Clock Regeneration,” European Conference on Optical Communication, ECOC 2001. Pages: 308-310. [28] Schmuck H., ’’Comparison of optical millimetre-wave system concepts with regard to chromatic dispersion,” Electron. Lett., vol. 31, pp. 1848-1849, 1995. [39] Shah V., Kim K.S., Morreale J.P., Chandrasekhar S., and Nichols V., “10 Gb/s transmission across 4 optical cross-connect nodes and 200 km fiber, “ European Conf. on Optical Comm., vol. 1, pp. 601-602,1998. [72] Shell M., Vaughn M. 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E., “Tunable compensation of dispersion-induced RF power degradation in multiple-channel SCM transmission by nonlinearly-chirped FBGs,” in Tech. Dig. Conf. on Lasers and Electro-Optics CLEO’99, paper CWK2, pp. 316-317, Baltimore MD, May 23-28,1999. [34] Sun H., Cardakli M.C., Feng K.-M., et al., “Tunable RF-power-fading compensation of multiple-channel double-sideband SCM transmission using a nonlinearly chirped FBG,” IEEE Photon. Technol. Lett., vol. 12 no. 5, pp. 546-548, 2000. [7] Tonguz O. K. and Jung H., "Personal Communications access networks using subcarrier multiplexed optical links," J. Lightwave Technol., vol. 14, pp. 1400-1409, June 1996. 97 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission. [23] Wood T. H., Feldman R. D., and Austin R. F., “Demonstration of a cost- effective broadband passive optical network system,” IEEE Photon. Technol. Lett., vol. 6, pp. 575-577,1994. [37] Xie Y., Lee S., Pan Z., Cai J.-X., Willner A.E, Grubsky V.,- Starodubov D.S., Salik E., and Feinberg J., "Tunable compensation of the dispersion slope mismatch in dispersion-managed systems using a sampled nonlinearly chirped FBG," IEEE Photon. Technol. Lett., vol. 12, Oct. 2000, pp. 1417-1419. [59] Yoo S.J.B., Chang G.K., Xin W., and Koza M.A., “Simultaneous Multi- Channel Conversion of Analog and Digital Signals by Polarization Independent Difference-Frequency-Generation,” Optical Fiber Comm. Conf., paper FB5,1999. [83] Yu C., Yu Q., Pan Z., Sahin A.B., Willner A.E., “Optical compensation of PMD-induced power fading for single sideband subcarrier-multiplexed systems,” OFC-2002, WQ5 ,2002 Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.

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Asset Metadata

Creator Sahin, Asaf (author)

Core Title Experimental demonstration of techniques to improve system performance in non-static microwave frequency analog and digital signal transmission over fiber -optic communication systems

Contributor Digitized by ProQuest (provenance)

School Graduate School

Degree Doctor of Philosophy

Degree Program Electrical Engineering

Publisher University of Southern California (original), University of Southern California. Libraries (digital)

Tag engineering, electronics and electrical,OAI-PMH Harvest

Language English

Advisor Willner, Alan E. (committee chair), Gagliardi, Robert (committee member), Rich, Daniel (committee member)

Permanent Link (DOI) https://doi.org/10.25549/usctheses-c16-650027

Unique identifier UC11335120

Identifier 3116779.pdf (filename),usctheses-c16-650027 (legacy record id)

Legacy Identifier 3116779.pdf

Dmrecord 650027

Document Type Dissertation

Rights Sahin, Asaf

Type texts

Source University of Southern California (contributing entity), University of Southern California Dissertations and Theses (collection)

Access Conditions The author retains rights to his/her dissertation, thesis or other graduate work according to U.S. copyright law. Electronic access is being provided by the USC Libraries in agreement with the au...

Repository Name University of Southern California Digital Library

Repository Location USC Digital Library, University of Southern California, University Park Campus, Los Angeles, California 90089, USA